DC/DC converter for switching a voltage level of DC voltage, and a control method for the DC/DC converter

ABSTRACT

The present invention suppresses influence of control due to the delay time in voltage switching between a plurality of different voltages. A DC/DC converter comprises: a main circuit including a switching circuit; a control unit that performs discrete control by discrete control toward a command value; and a switching signal generation unit that generates a switching signal that drives the switching circuit. The control unit calculates the pulse width ΔT(k) of the switching signal having the delay time Td as a parameter as an operation amount of discrete control performed every control period Ts. The switching signal generation unit generates a switching signal based on the pulse width ΔT(k) obtained by the calculation in the control unit. The main circuit switches the voltage level by driving the switching circuit based on the switching signal of the switching signal generation unit.

TECHNICAL FIELD

The present invention relates to a DC/DC converter for switching avoltage level of a DC voltage, and a control method for the DC/DCconverter.

BACKGROUND ART

(High/Low Pulse Operation)

In recent years, for example, high-frequency power (RF output) has beenused in the field of plasma application, which power is generated by anON/OFF pulse operation for turning ON/OFF in a cycle of several tens ofHz to several tens of kHz or a High/Low pulse operation for varying anamplitude of RF power at high speed.

These pulse operations are said to be effective for suppressing abnormaldischarge occurring due to particles produced during film formation andfor microfabrication and others by using low-temperature plasma.

The ON/OFF pulse operation is an operation mode of supplyingintermittent high frequency power (RF output) to load. In this operationmode, plasma may be extinguished in an OFF period where the power is notsupplied to the load. As a consequence, once the plasma is extinguished,the RF output will have a mismatch with a plasma impedance.

By contrast, the High/Low pulse operation is an operation mode ofperiodically varying continuous high frequency power, which does notintermit at all times, to the load by dividing different two levels of ahigh level and a low level, thereby supplying power at a level differentfrom the high level, instead of utilizing the OFF period of the ON/OFFpulse operation. For example, in the power supplying to the plasma, acontinuous output is produced between power on a high side that isnecessary to form a thin film and power on a low side that is necessaryto maintain plasma discharge to thereby prevent the plasma from beingextinguished and maintain constant stable plasma discharge.

(DC/DC Converter)

For the generation of high-frequency power, there is a method forperforming the High/Low pulse operation by controlling a DC/DC convertersection in high-frequency power generation.

Since it is required to make a fast transition between two differentvoltage levels to control the DC/DC converter section, a frequencylimitation of the High/Low pulse operation is dependent on controlresponsivity of the DC/DC converter. Thus, in order to make the fasttransition between the voltage levels, it is necessary to change thevoltage quickly in the DC/DC converter and control the voltage stably.

As a control method of the DC/DC converter, a PI control is well known.The PI control is a classic way of conducting the control in which adifference between a command value and a detection value is proportionedand integrated to calculate a manipulated variable. As an example, thereis a PI control adopting a double closed loop control system comprisinga minor loop using a capacitance current and a major loop using anoutput voltage. The PI control of the closed loop control type isclassical, and the control responses of the major loop and the minorloop have the following limitations.

1) Since the minor loop is affected by, such as, dead time, the maximumcontrol response thereof is a frequency of about 1/10 of a switchingfrequency.

2) Due to the prevention of interference with the minor loop, themaximum control response of the major loop is a frequency of about 1/10of the control response of the minor loop.

Thus, the maximum control response of the major loop is a frequency ofabout 1/100 of the switching frequency. Due to this limitation in thecontrol response, when the High/Low pulse operation is conducted at afrequency of 10 kHz or more, the switching frequency will exceed 1 MHzto thereby cause control complication, and the control response of theclosed loop control will exceed the limitation. Accordingly, it isdifficult in the PI control to achieve a stable High/Low pulse operationthat can gain fast rise time and fall time.

(Discrete Control)

As a control method for a DC/DC converter with high responsivity, thereis a discrete control. FIG. 20 shows a diagram of PI control anddiscrete control. PI control illustrated in FIG. 20(a) calculates amanipulated variable by detecting an error between an output and acommand value, so as to gradually follow the command value according toa control response frequency.

By contrast, the discrete control shown in FIG. 20(b) uses a model of amain circuit of the DC/DC converter and a detection value to calculate amanipulated variable required for matching a control value with adesired value after one sample. The manipulated variable is then fed tothe main circuit to carry out non-linear control for matching a commandvalue with the control value at a next sample point.

The discrete control computes a pulse width ΔT(k) for each samplingperiod so that a control value of the (ks+1)-th sampling period becomesequal to the desired value for a state equation obtained by expanding acircuit state with an input and an output as state variables by adiscrete model, and the output is then controlled by a switchingoperation according to the obtained pulse width ΔT(k).

In the discrete control, a switching frequency remains the maximumcontrol response in an ideal state. In this case, the manipulatedvariable of the discrete control is determined by using a relationalexpression of the modeled main circuit.

Non-Patent Literature 1 suggests control using a voltage detection valueonly. In addition, Non-Patent Literatures 2 to 4 teach control ofestimation of a delay to implement compensation. Moreover, Non-PatentLiterature 5 refers to an influence on the stability by a delay timecaused by averaging in digital control.

With respect to the control by the High/Low pulse operation,ILrefcontrol using an inductance current iL as detection value has beensuggested (Non-Patent Literature 6). The ILref control is output controlperformed by using the inductance current as desired value andconsidering an output current Iout as disturbance. Non-Patent Literature7 teaches that a transition by 108V from Low 12V to High 120V isachieved in 518 μs at a switching frequency of 200 kHz.

CITATION LIST Non-Patent Literature

-   Non-Patent Literature 1: A. Kawamura, T. Haneyoshi, and R. G. Hoft,    “Deadbeat Controlled PWM Inverter with Parameter Estimation Using    Only Voltage Sensor”, IEEE transactions on Power Electronics, Vol.    3, Issue 2, pp. 118-125 (1988)-   Non-Patent Literature 2: C. Li, S. Shen, M. Guan, J. Lu, and J.    Zhang, “A Delay-compensated Deadbeat Current Controller for AC    Electronic Load”, In Proceeding of the 25th Chinese Control    Conference, CCC 2006, pp. 1981-1985 (2006)-   Non-Patent Literature 3: K. Hung, C. Chang, and L. Chen, “Analysis    and Implementation of a Delay-compensated Deadbeat Current    Controller for Solar Inverters”, In Proceeding of Circuits, Devices    and Systems, Vol. 148, pp. 279-286 (2001)-   Non-Patent Literature 4: T. Nussbaumer, M. L. Heldwein, G.    Gong, S. D. Round, and J. W. Kolar, “Comparison of Prediction    Techniques to Compensate Time Delays Caused by Digital Control of a    Three-Phase Buck-Type PWM Rectifier System”, IEEE Transactions on    Industrial Electronics, Vol. 55, Issue 2, pp. 791-799 (2008)-   Non-Patent Literature 5: J. Chen, A. Prodic, R. W. Erickson, and D.    Maksimovic, “Predictive Digital Current Programmed Control”, IEEE    Transactions on Power Electronics, Vol. 18, Issue 1, pp. 411-419    (2003)-   Non-Patent Literature 6: S. Mizushima, A. Kawamura, I.    Yuzurihara, A. Takayanagi, and R. Ohma, “DC Converter Control Using    Deadbeat Control of High Switching Frequency for Two-type Operation    Modes”, In Proceeding of the 40th Annual Conference of the IEEE,    IECON 2014, Vol. 1, pp. 5029-5034 (2014)-   Non-Patent Literature 7: S. Mizushima, H. Adachi, A. Kawamura, I.    Yuzurihara, and R. Ohma, “High/Low Pulse Generation of Deadbeat    Based High Power DC-DC converter with Very Short Rise Time”, In    Proceeding of the 8th International Power Electronics and Motion    Control Conference of the IEEE, IPEMC-ECCE Asia 2016, Vol. 1, pp.    609-615 (2016)

SUMMARY OF INVENTION Problems to be Solved by the Invention

In the High/Low pulse operation of the DC/DC converter, if a transitiontime taken for rising during the transition from the low side power tothe high side power and a transition time taken for falling during thetransition from the high side power to the low side power are slow,unstable plasma is generated in a transition period that leads to theformation of an uneven thin film. Thus, it is required to enhance thespeed of rising and falling to shorten the transition times.

When the switching frequency is increased to achieve high-speedresponse, a delay time developed between a main circuit and a controlunit has a measurable influence on the acquisition of voltage andcurrent detection values. The delay time is an acquisition delay in adetector, a computation delay developed when computing manipulatedvariables for discrete control or others and a response delay in aswitching device of a DC/DC converter, by way of example.

For instance, when high-frequency pulse power is generated bycontrolling a High/Low pulse operation of a DC/DC converter to performplasma control with the generated high-frequency pulse power, transitiontime having a short μs scale, which is time for plasma iongeneration/disappearance, is determined in voltage changes on the risingedge and falling edge of a high-frequency pulse signal. A DC currentsensor detecting an inductance current in the DC/DC converter has anunignorable delay time of about 1 μs, even if the sensor is a high-speeddevice. The delay time may cause an error between an object to becontrolled and a control model, and the error may develop a problem inthe accuracy of the discrete control, resulting in controlledoscillation.

Thus, it is required to perform the high-speed control in the voltageswitching between the plural different voltages, while suppressing theinfluence of the control due to the delay time.

The present invention aims to solve the existing problem and suppressthe influence of the control due to the delay time during the voltageswitching between the plural different voltages.

Means for Solving the Problem

With respect to the problem of the inevitable delay time, such ascomputation delay in the control operation, the present inventionconducts compensation by modeling the main circuit in the discretecontrol. The discrete control of the invention is for defining a pulsewidth ΔT that enables to obtain an output according to a command valueat a point after n-th sample from the present point, wherein n may be anarbitrary integer, and if n is “1”, the control will be conducted at thepoint after one sample.

The compensation by modeling the main circuit is carried out by derivingthe manipulated variable for the discrete control which takes intoaccount the presence of a delay time Td. As a manipulated variable forthe discrete control taking into account the delay time Td, a pulsewidth ΔT(k) of the discrete control to which a parameter of the delaytime Td is added. It can realize the control of a DC/DC converter withhigh-speed responsivity. Moreover, the application of this technique cansuppress the influence of the control due to the delay time during thevoltage switching between the plural different voltages, therebyenabling to perform fast voltage switching. When the present inventionis applied to the High/Low pulse operation, the voltage switching isperformed between a high voltage level and a low voltage level.

Since there are many AC current sensors with response capability of 10MHz or more, the DC current sensor is replaced by an AC current sensorwhile using a capacitance current which can be detected as a detectionvalue of the discrete control by the AC current sensor.

The present invention has aspects of a DC/DC converter and a controlmethod for the DC/DC converter.

(DC/DC Converter)

A DC/DC converter comprises a main circuit including a switchingcircuit, a control unit for performing discrete control toward a commandvalue, and a switching signal generator for generating a switchingsignal that drives the switching circuit.

The control unit computes a pulse width ΔT(k) of a switching signalhaving a delay time Td as a parameter for a manipulated variable for thediscrete control conducted for every control cycle Ts. The switchingsignal generator generates the switching signal based on the pulse widthΔT(k) obtained by the computation in the control unit. The main circuitswitches a voltage level by driving the switching circuit based on theswitching signal of the switching signal generator.

In the configuration that the main circuit comprises the switchingcircuit and an LC circuit composed of a series-parallel circuit of aninductance LA and a capacitance C, the pulse widthsΔT(k) of a cycle (k)of the main circuit in the discrete control by the control unit, havinga voltage component across the capacitance C in a control cycle (ks) andan input voltage vo in the control cycle (ks) in the LC circuit, and apulse width ΔT(k−1) of a cycle (k−1) of the main circuit, are term oftime.

For example, the voltage component across the capacitance C when theinductance current is used as detection value is((Ts+Td)²/2C)·iLA-ave(ks)/Vin and ((Ts+Td)²/2C)·iR(ks)/Vin, and thevoltage component across the capacitance C when the capacitance currentis used as detection value is ((Ts+Td)²/2C)·iC-ave(ks)/Vin. Both of thecomponents are the term of time of (Ts+Td)². The input voltage vo is((Ts+Td)·vo(ks))/Vin, which is the term of time of (Ts+Td). In additionto that, the pulse width ΔT(k−1) is the term of time having acoefficient of (Td/Ts). A formula expressing each pulse width ΔT(k) willbe described below.

The term of time of each voltage component has an additional value(Ts+Td) as a parameter, which is obtained by adding the delay time Td tothe control cycle Ts. The term of time of the pulse width ΔT(k−1) has acoefficient of (Td/Ts) obtained by dividing the delay time Td by thecontrol cycle Ts. The DC/DC converter of the present invention has theparameter of the delay time Td in the pulse width ΔT(k), therebysuppressing the influence of the control due to the delay time.

The present invention can adopt a single-phase or multi-phaseinterleaving control. The inductance value of the pulse width ΔT(k) inthe multi-phase interleaving control of n-phase (wherein n is an integerof 2 or more) is 1/n of the inductance value of the pulse width ΔT(k) inthe single-phase interleaving control.

(Single-Phase)

In the single-phase discrete control, the main circuit comprises asingle-phase switching circuit and a single-phase LC circuit composed ofa series-parallel circuit of an inductance LA and a capacitance C.

The control unit performs the computation to obtain as a manipulatedvariable for the single-phase discrete control the pulse width ΔT(k) ofthe switching signal with the parameter of (Ts+Td) obtained by addingthe delay time Td to the control cycle Ts.

In the single-phase discrete control, the pulse width ΔT(k) as amanipulated variable of the discrete control is expressed by thefollowing formula when an inductance current iLA is used as detectionvalue.

$\begin{matrix}{{{\Delta T}(k)} = {\frac{{L_{A}{i_{LA}\left( {k + 1} \right)}} - {\left\{ {L_{A} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{{LA} - {ave}}\left( k_{s} \right)}}}{V_{in}} + \frac{{\left( {T_{s} + T_{d}} \right){v_{o}\left( k_{s} \right)}} - \frac{\left( {T_{s} + T_{d}} \right)^{2}{i_{R}\left( k_{s} \right)}}{2C}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}} & \left( {{Formula}1} \right)\end{matrix}$

The pulse width ΔT(k), when a capacitance current iC is used asdetection value, is expressed by the following formula.

$\begin{matrix}{{{\Delta T}(k)} = {\frac{{LI}_{Cref} - {\left\{ {L - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{C - {ave}}\left( k_{s} \right)}}}{V_{in}} + \frac{\left( {T_{s} + T_{d}} \right){v_{o}\left( k_{s} \right)}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}} & \left( {{Formula}2} \right)\end{matrix}$

The pulse width ΔT(k) expressed by the above formula can be obtainedwhen an average value is used as detection value. In this formula,iLA-ave(ks) is an average value of the inductance current iLA,iC-ave(ks) is an average value of the capacitance current iC, andv_(o)(ks) is an average value of a detected output value vo. For theaverage current iLA-ave of the inductance current and the capacitancecurrent iC-ave(ks) at the point ks, average values in a period of[ks−1−ks] is used, which is one cycle of the switching, and for anaverage value input of [k−k+1] derived at the point ks, a valueVin·Duty(=ΔT(k)/Ts) is used.

In the case of using the detection value at each point in the controlcycle, the expression of the discrete control can be obtained by usingthe detection value at each point in the control cycle, instead of theaverage values presented in the above formula.

The above-described parameter width ΔT(k) is an approximate expressionin which a general solution x(t) of a state equation is approximated bya quadratic expansion expression, and includes a quadratic term for(Ts+Td) that includes the control cycle Ts and the delay time Td. Theapproximation of the general solution x(t) of the state equation is notlimited to the approximation according to the quadratic expansionexpression, and thus a higher-order expansion expression may be used forapproximation that can increase the degree of approximation of the widthΔT(k).

The switching signal generator generates a single-phase switching signalbased on the manipulated variable (pulse width ΔT(k)) to drive theswitching circuit of the main circuit.

(Multi-Phase: n-Phase)

In the multi-phase discrete control, the main circuit comprises ann-phase LC circuit composed of a series-parallel circuit of aninductance L and a capacitance C of each phase which are connected inparallel.

The control unit performs the computation to acquire as a manipulatedvariable for the n-phase discrete control the pulse width ΔT(k) of theswitching signal having the parameter of (Ts+Td) obtained by adding thedelay time Td of the control cycle Ts.

The pulse width ΔT(k) as a manipulated variable for the discrete controlby the n-phase interleaving in the control unit is expressed by thefollowing formula when the inductance current iLA is used as detectionvalue.

$\begin{matrix}{{{\Delta T}(k)} = {\frac{{\frac{L}{n}{i_{L}\left( {k + 1} \right)}} - {\left\{ {\frac{L}{n} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{L - {ave}}\left( k_{s} \right)}}}{V_{in}} + \frac{{\left( {T_{s} + T_{d}} \right){v_{o}\left( k_{s} \right)}} - {\frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}{i_{R}\left( k_{s} \right)}}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}} & \left( {{Formula}3} \right)\end{matrix}$

When the capacitance current iC is used as detection value, the pulsewidth ΔT(k) is expressed by the following formula.

$\begin{matrix}{{{\Delta T}(k)} = {\frac{{\frac{L}{n}I_{Cref}} - {\left\{ {\frac{L}{n} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{C - {ave}}\left( k_{s} \right)}}}{V_{in}} + \frac{\left( {T_{s} + T_{d}} \right){v_{o}\left( k_{s} \right)}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}} & \left( {{Formula}4} \right)\end{matrix}$

The above-described pulse width ΔT(k) is the case of using the averagevalues of the inductance current and the capacitance current as currentdetection values. In the above formula,

Vin(k) is an input voltage;

vo(ks) is an output voltage at the point ks in the control cycle;

iL(k) is a combined current obtained from the inductance currents at thepoint k in the cycle of the main circuit;

iL-ave(ks) is an average value of the combined current of the inductancecurrents between the point ks−1 and point ks in the control cycle;

iC(k) is a combined current obtained from the capacitance currents atthe point k in the cycle of the main circuit;

iC-ave(ks) is an average value of the combined current of thecapacitance currents between the point ks−1 and the point ks in thecontrol cycle;

iR(ks) is a load current at the point ks in the control cycle;

ICref is a command value of the capacitance current;

L is the inductance of the LC circuit; and

C is the capacitance of the LC circuit.

The parameter width ΔT(k) in the case of where the n-phase isthree-phase is expressed by the following formula when the inductancecurrent iLA is used as detection value.

$\begin{matrix}{{{\Delta T}(k)} = {\frac{{\frac{L}{3}{i_{L}\left( {k + 1} \right)}} - {\left\{ {\frac{L}{3} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{L - {ave}}\left( k_{s} \right)}}}{V_{in}} + \frac{{\left( {T_{s} + T_{d}} \right){v_{o}\left( k_{s} \right)}} - {\frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}{i_{R}\left( k_{s} \right)}}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}} & \left( {{Formula}5} \right)\end{matrix}$

The pulse width ΔT(k) when the capacitance current iC is used asdetection value is expressed by the following formula.

$\begin{matrix}{{{\Delta T}(k)} = {\frac{{\frac{L}{3}I_{Cref}} - {\left\{ {\frac{L}{3} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{C - {ave}}\left( k_{s} \right)}}}{V_{in}} + \frac{\left( {T_{s} + T_{d}} \right){v_{o}\left( k_{s} \right)}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}} & \left( {{Formula}6} \right)\end{matrix}$

The switching signal generator generates an n-phase switching signal onthe basis of the manipulated variable (pulse width ΔT(k)) so as to drivethe switching circuit of the main circuit.

(Three-Mode Control for Voltage Level Switching)

The DC/DC converter of the present invention has an aspect of three-modecontrol in switching a DC input to different voltage levels.

The control unit repeats the following three modes to convert the DCinput into high-frequency pulse outputs at different voltage levels:

a first mode that conducts constant-current control to implement atransition period for the transition between a first power level and asecond power level;

a third mode that conducts constant-voltage control to implement aretention period for retaining each voltage at the first power level andthe second power level; and

a second mode that conducts the constant-voltage control to implement abuffer period to shift from the transition period to the retentionperiod.

By changing the voltage level for the switching, the voltage levels canbe switched between the plural voltage levels (multiple levels).

With respect to the manipulated variable (pulse width ΔT(k)) for thediscrete control by the n-phase interleaving in each mode:

the manipulated variable (pulse width ΔT(k)) for the discrete control inthe first mode is expressed by the following formula;

$\begin{matrix}{{{\Delta T}(k)} = {\frac{{\frac{L}{n}I_{Cref}} - {\left\{ {\frac{L}{n} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{C - {ave}}\left( k_{s} \right)}}}{V_{in}} + \frac{\left( {T_{s} + T_{d}} \right){v_{odet}\left( k_{s} \right)}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}} & \left( {{Formula}7} \right)\end{matrix}$

the manipulated variable (pulse width ΔT(k)) for the discrete control inthe second mode is expressed by the following formula;

$\begin{matrix}{{{\Delta T}(k)} = {\frac{\begin{matrix}{{\frac{L}{n}A_{H1}\left\{ {V_{ret} - {V_{odet}\left( k_{s} \right)}} \right\}} -} \\{\left\{ {\frac{L}{3} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{C - {ave}}\left( k_{s} \right)}}\end{matrix}}{V_{in}} + \frac{\left( {T_{s} + T_{d}} \right){v_{odet}\left( k_{s} \right)}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}} & \left( {{Formula}8} \right)\end{matrix}$

the manipulated variable (pulse width ΔT(k)) for the discrete control inthe third mode is expressed by the following formula.

$\begin{matrix}{{{\Delta T}(k)} = {\frac{{\left( {T_{s} + T_{d}} \right)V_{ref}} - {\left\{ {\frac{L}{n} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{C - {ave}}\left( k_{s} \right)}}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}} & \left( {{Formula}9} \right)\end{matrix}$

In the above formulas,

Vin(k) is an input voltage,

Vref is a voltage command value,

vodet(ks) is an output voltage detected value obtained from initialvalues vo(ks) and vo(ks−1),

ICref is a command value of the capacitance current,

iC-ave(ks) is an average value of a triangular ripple current obtainedfrom the end of multiplication of n-phase and can be obtained by theaverage value of the capacitance current at the point ks in the controlcycle,

L is the inductance of the LC circuit, and

C is the capacitance of the LC circuit.

The switching signal generator generates an n-phase switching signalbased on the manipulated variable.

(Control Method for DC/DC Converter)

A control method for a DC/DC converter comprising a main circuitincluding a switching circuit, a control unit for performing discretecontrol toward a command value, and a switching signal generator forgenerating a switching signal that drives the switching circuit, whereinthe control unit computes a pulse width ΔT(k) of a switching signalhaving a delay time Td as a parameter for a manipulated variable of thediscrete control conducted for every control cycle Ts, the switchingsignal generator generates the switching signal based on the pulse widthΔT(k) thus obtained, and the main circuit drives the switching circuitbased on the switching signal. The discrete control may have an aspectof a single-phase or multi-phase interleaving control.

Effects of the Invention

The DC/DC converter of the present invention uses the pulse width ΔT(k)of the discrete control with the parameter of delay time Td asmanipulated variable for the discrete control that takes intoconsideration the delay time Td so as to conduct the modeling of themain circuit in the discrete control, thereby suppressing the influenceon the control due to the delay time in the voltage switching betweenthe different voltage levels. Consequently, it allows fast voltageswitching, thereby suppressing the influence of an inevitable delay timein the control, such as computation delay.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a diagram illustrating a schematic configuration example of aDC/DC converter according to the present invention;

FIG. 2 is a schematic view showing circuitry of a single-phase step-downDC/DC converter in the DC/DC converter according to the presentinvention;

FIG. 3 is a diagram illustrating a relationship of delay times Tdbetween a control circuit (controller) and a main circuit in the DC/DCconverter according to the present invention;

FIG. 4 is a diagram illustrating a case where the delay times Td do notoccur in a cycle relationship between the control circuit (controller)and the main circuit;

FIG. 5 is a diagram illustrating a case where the delay times Td occurin the cycle relationship between the control circuit (controller) andthe main circuit;

FIG. 6 is a diagram illustrating a case where the delay times Td occurin the cycle relationship between the control circuit (controller) andthe main circuit in the present invention;

FIG. 7 is a diagram illustrating a relationship between a control periodTs and the delay time Td as well as an integration period;

FIG. 8 is a diagram illustrating an example of applying a three-phaseinterleaving system;

FIG. 9 is a diagram illustrating a schematic configuration of step-downDC/DC converter by the three-phase interleaving system;

FIG. 10 is a diagram illustrating an equivalent circuit of the step-downDC/DC converter of FIG. 9 ;

FIG. 11 is a diagram illustrating an equivalent circuit which is onephase of bidirectional step-down chopper circuit by the three-phaseinterleaving system;

FIG. 12 is a diagram illustrating an average period for obtaining anaverage current;

FIG. 13 is a diagram illustrating a controlling form by a combination ofconstant-voltage control and constant-current control;

FIG. 14 is a diagram illustrating each mode in a High/Low pulseoperation of discrete control according to the present invention;

FIG. 15 is a diagram illustrating a controlling form of each mode in thediscrete control and parameters according to the present invention;

FIG. 16 is a diagram illustrating a controlling form of each mode in thediscrete control based on three phases according to the presentinvention;

FIG. 17 is a diagram illustrating signal states in the discrete controlby the mode I, mode II and mode III according to the present invention;

FIG. 18 is a flowchart illustrating a mode transition when shifting froma low power side to a high power side;

FIG. 19 is a diagram illustrating examples of the application of theDC/DC converter according to the present invention to a DC power sourcedevice and an AC power source device; and

FIG. 20 is a schematic view illustrating PI control and the discretecontrol.

BEST MODE FOR CARRYING OUT THE INVENTION

A description will now be made on a DC/DC converter and a control methodfor the DC/DC converter according to the present invention by referringto the accompanying drawings. In the following description, FIG. 1 isreferred to illustrate a schematic configuration example of the DC/DCconverter of the present invention, FIGS. 2 to 7 are referred toillustrate discrete control according to the present invention bysingle-phase, FIGS. 8 to 12 are referred to illustrate the discretecontrol according to the present invention by multi-phase, and FIGS. 13to 18 are referred to illustrate each mode in the discrete controlaccording to the present invention.

(Schematic Configuration of the DC/DC Converter of the PresentInvention)

With reference to FIG. 1 , a schematic configuration of the DC/DCconverter of the present invention will be described. A DC/DC converter1 of the present invention comprises a main circuit (LC chopper circuit)2 configured to use an input voltage Vin as an input to output adetected output voltage vo and a load current iR, a switching signalgenerator 5 configured to generate a switching signal for controlling anON/OFF operation of a switching device of the main circuit 2, and acontrol unit 6 configured to input detection signals from the maincircuit 2 and a load 7 to compute a pulse width ΔT(k), and then outputthe computed pulse width ΔT(k) to the switching signal generator 5.

The LC chopper circuit of the main circuit 2 includes an LC circuit 4formed by connecting an inductance L and a capacitance C inseries-parallel, and a switching circuit 3 configured to conductmulti-phase switching control on the input voltage Vin to thereby supplythe LC circuit 4 with an inductance current iL thus produced.

The control unit 6 computes the pulse width ΔT(k) of the switchingsignal for controlling the ON/OFF operation of the switching device ofthe switching circuit 3. The pulse width ΔT(k) is for determining a timewidth in an ON state of the switching device in one cycle of switching.The control unit 6 controls power supplied to the load 7 through the LCcircuit 4 based on the length of the pulse width ΔT(k). If the timewidth of the switching cycle is defined as Ts, the control unit 6 maycompute a duty ratio Duty(=ΔT(k)/Ts) of the pulse width ΔT(k) withrespect to the time width T to thereby perform the control based on theduty ratio. The control unit 6 implements the controlling forms ofvoltage control, current control and power control according to the formof a designated value.

As shown in FIG. 20(b), the control unit 6 uses an output of (ks+1)-thsampling cycle as a control value to compute the pulse width ΔT(k) forevery sampling cycle such that the control value becomes equal to acommand value which is a desired value, and thus conduct discretecontrol to control the switching operation on the basis of the computedpulse width ΔT(k). The control unit 6 carries out constant-currentcontrol in the discrete control in a predetermined cycle based on acontrol current including a phase current in the main circuit 2, so asto conduct the computation for every sampling cycle Ts on the pulsewidth ΔT(k) of the switching signal that actuates a switching device,not illustrated, of the switching circuit 3 of the main circuit 2. Inthe discrete control by multi-phase interleaving, a combined currentobtained by combining a phase current value in each phase is used toproduce a control signal. In this context, the sampling cycle isutilized as switching cycle.

The control unit 6 determines the pulse width ΔT(k) computed by carryingout the constant-current control on the control current including thecombined current as the pulse width ΔT(k) of each phase current. Bycarrying out the constant-current control on the control current, a stepresponse becomes a step response for the current, not for a voltage,thereby suppressing a secondary oscillating voltage of an outputvoltage.

The switching signal generator 5 of the present invention generates aswitching signal for each phase with the pulse width ΔT(k) computed bythe control unit 6 being set as the pulse width ΔT(k) for each phase. Inthe computation of the pulse width ΔT(k), the pulse width ΔT(k) iscomputed based on the control current including the combined currentthat is obtained by combining the phase current values. In thiscomputation, since the control current is based on the combined currentof the phase current values, limitations caused by overlapping betweenthe pulse widths ΔT(k) of the respective phases can be eliminated,thereby enabling to obtain the pulse width ΔT(k) where the pulse widthsΔT of the phases are allowed to overlap one another.

(Discrete Control)

The discrete control of the present invention is for controlling thepulse width ΔT so as to obtain an output according to a command valuefrom the present point to a point after n-th sample. In this context, nmay be an arbitrary integer, and if n is “1”, the control will beconducted at the point after one sample.

There is control of a feedback gain that is known for setting statevariables of currents and voltages after n-th sample. This control isso-called deadbeat control. The discrete control of the presentinvention is similar to the deadbeat control in terms of performing thecontrol for a predetermined value after the n-th sample. However,instead of obtaining the feedback gain, the discrete control determinesthe pulse width ΔT that define the power in each control cycle asmanipulated variables required for the discrete control.

In the discrete control, in order to derive the manipulated variablerequired of a controlled variable to follow the command value after onesample, modeling is performed by using a state equation of the maincircuit to be controlled. For distinguishing single-phase alternatingcurrent and multi-phase alternating current of a commercial AC signal inthe control of the DC/DC converter of the present invention, controlperformed with a series of control signals of a predetermined cycle assingle phase is referred to as single-phase control, and controlperformed with a plurality of series of control signals of thepredetermined cycle with their phases shifted from one another isreferred to as multi-phase control. The DC/DC converter of the presentinvention can be applied to the single-phase as well as the multi-phase.Now, the single-phase control will be described first, and then themulti-phase control will be described. In regard of the multi-phase(n-phase) control, the description will be made about three-phase.

<State Equation of Main Circuit>

FIG. 2 illustrates an example of circuitry in a single-phase step-downDC/DC converter. The DC/DC converter comprises a switching circuitincluding a switching device S1A connected in series and a switchingdevice S2A connected in parallel between an input voltage Vin and a loadRL, and an LC circuit including an inductance LA connected in serieswith respect to the switching circuit and the load and a capacitance Cconnected in parallel with respect to the switching circuit and theload.

In this DC/DC converter, when input voltages applied across the LCcircuit by the switching devices S1A and S2A are defined as u1(t), acircuit equation of the LC circuit can be expressed by the followingformula.

$\begin{matrix}\left( {{Formula}{}10} \right) & \end{matrix}$ $\begin{matrix}\left. \begin{matrix}{{u_{1}(t)} = {{L_{A}\frac{{di}_{LA}}{d_{t}}} + V_{o}}} \\{{C\frac{{dV}_{o}}{{dt}_{o}}} - i_{LA} - {\frac{1}{R}V_{o}}}\end{matrix} \right\} & (1)\end{matrix}$

On the basis of the above circuit equation, the following state equationcan be obtained.(Formula 11){dot over (x)}(t)=Ax(t)+Bu(t)  (2)

In this regard, x(t), A, B, u(t) are as below.

$\begin{matrix}\left( {{Formula}12} \right) & \end{matrix}$ $\begin{matrix}\left. \begin{matrix}\begin{matrix}\begin{matrix}{{x(t)} = \begin{bmatrix}{i_{LA}(t)} & {v_{o}(t)}\end{bmatrix}^{T}} \\{{u(t)} = {u_{1}(t)}}\end{matrix} \\{A = \begin{bmatrix}0 & {{- 1}/L_{A}} \\{1/C} & {{- 1}/{CR}}\end{bmatrix}}\end{matrix} \\{B = {\frac{1}{L_{A}}\begin{bmatrix}1 \\0\end{bmatrix}}}\end{matrix} \right\} & (3)\end{matrix}$

<Derivation of Formula for Discrete Control Considering Delay Time>

Next, the state equation of the main circuit is used to derive a formulafor discrete control. The general solutions obtained by the above stateequations (2), (3) can be divided into sections, where an input u1(τ) isconstant, and are expressed by the following Formula 4.(Formula 13)x(t)=e ^(At) x(0)+∫₀ ^(t) e ^(A(t-σ)) Bu ₁(σ)dσ  (4)

The general solution of Formula 4 is used to derive a relationalexpression between the command value and the manipulated variable in thediscrete control. There are delay times between the control conducted bythe control unit (controller) and the operation in the main circuit dueto acquisition delay occurring during acquisition of a detection valueof the voltage or current in the detector, calculation delay occurringduring calculation of the manipulated variable from the detection valuein the control unit, operation delay occurring in the switching devicein the main circuit from receiving a gate signal till going intooperation. The delay time causes an error between a real circuit to becontrolled and a control model, leading to a problem in the accuracy ofthe discrete control that may cause a controlled oscillation.

In order to take into consideration the delay time, the formula for thediscrete control, into which the term of the delay time is introduced,is derived so as to derive the manipulated variable for controlling theoutput of the main circuit to the command value. The control circuit(controller) generates the pulse width ΔT(k) for controlling the outputby turning on and off the input voltage as manipulated variable forcontrolling the switching of the main circuit based on the formula forthe discrete control. The delay time is represented below as Td.

The switching operation of the switching circuit is implemented by agate signal output from the control unit. The switching operation can beconducted with a single-phase gate signal as well as a multi-phase gatesignal based on multiple phases (n-phase), and the switching operationwith the multi-phase gate signal is multi-phase interleave.

Now, a description will be made first about the formula for discretecontrol in single-phase, and then about the formula for discrete controlin multi-phase (n-phase). As to the multi-phase (n-phase) control, thedescription will refer to the three-phase control.

(Formula for Single-Phase Discrete Control)

Regarding the case of implementing the switching operation bysingle-phase, the derivation of the formula for discrete control inconsideration the delay time Td will be described below.

The delay time Td varies in terms of time axis depending on theabove-mentioned various factors. FIG. 3 illustrates a case assuming thatthe delay time Td is within one cycle of a sampling cycle Ts. In FIG. 3, k represents a control cycle in the main circuit and ks represents acontrol cycle (sampling cycle) of a control unit (controller).

The discrete control conducts the control after one cycle of the controlcycle k of the main circuit so that an output of the main circuitfollows a command value. Thus, at the point ks in the control cycle ofthe control unit (controller), the general solution of the stateequation up to the point (k+1), which is the end of one cycle in thecontrol cycle of the main circuit, is computed. In order to taking intoconsideration the delay time Td, the term of the delay time Td isintroduced to the general solution to thereby obtain the pulse widthΔT(k) of the manipulated variable of the discrete control.

The control unit (controller) computes the manipulated variable at thepoint of ks for matching the controlled variable with the command valueat the point (k+1) in the control cycle of the main circuit, and thenobtain the pulse width ΔT(k) based on the manipulated variable. Theswitching circuit uses a gate signal formed on the basis of the obtainedpulse width ΔT(k) to open and close the switching device of the maincircuit.

In the state equations (2) and (3), the general solution x(t) of eachstate equation includes an inductance current iLA(t) and a detectedoutput voltage vo(t).

In the computation of the manipulated variable, a value at each point inthe control cycle is used for the inductance current iLA(t) in theequation. As the inductance current iLA(t) contains a ripple componentat the time of switching, a current value varies in one cycle of thecycle k of the main circuit in the single-phase control. Thus, inaddition to use the value at each point in the control cycle, adetection value of the inductance current iLA(t) may be obtained from anaverage value in one cycle of the cycle k of the main circuit so as tosuppress the influence of fluctuations in the current value. Since theripple component of the inductance current iLA(t) does not affect arising voltage and a falling voltage of the detected output voltage vo,the fluctuation in the detected output voltage vo has no influence onthe average value.

As to the input voltage u1(t), the value at each point in the controlcycle is used in the equation. The input voltage u1(t) of the maincircuit has a pulsed input waveform depending on the gate signal. Thus,in addition to use the value at each point in the control cycle, adetection value of the input voltage may also be obtained from anaverage value in one sampling cycle of the control cycle ks, in order toavoid fluctuations in the detected voltage value due to the differencein the sampling points. Since the input voltage u1(t) is output with atime width determined by a duty of the pulse width ΔT(t) of the gatesignal for two values of the input voltage value Vin and a zero voltage,the average value of one sampling cycle of the control cycle ks iscalculated as input voltage value Vin·Duty. In this context, the dutydetermined at the point (ks−1) is presented by a ratio ΔT(k−1)/Ts of atime width ΔT(k−1) of the gate signal with respect to the time width Tsfor one cycle, and the duty of ΔT(k) derived from the detection value atthe point ks is represented by a ratio ΔT(k)/Ts of the time width ΔT(k)of the gate signal with respect to the time width Ts for one cycle.

As described above, it is not necessary to use the average values forthe inductance current iLA(t) of the detected current and the inputvoltage u1(t) depending on the number of the phases of the interleaveand the delay time, so that the detection value at each point in thecontrol cycle can be used.

By using the general solution of Formula 4, the pulse width ΔT(k) as themanipulated variable of the discrete control considering the delay timeTd is expressed by the following Formula 5.

$\begin{matrix}\left( {{Formula}14} \right) & \end{matrix}$ $\begin{matrix}{{{\Delta T}(k)} = {\frac{{L_{A}{i_{LA}\left( {k + 1} \right)}} - {\left\{ {L_{A} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{{LA} - {ave}}\left( k_{s} \right)}}}{V_{in}} + \frac{{\left( {T_{s} + T_{d}} \right){v_{o}\left( k_{s} \right)}} - {\frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}{i_{R}\left( k_{s} \right)}}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}} & (5)\end{matrix}$

In Formula 5, the pulse width ΔT(k) in the cycle (k) of the main circuithas in the LC circuit a voltage component LA·iLA(k+1) in the inductanceLA due to iLA(k+1) in the next cycle (k+1) of the main circuit, and avoltage component (LA−(Ts+Td)²/2C)·iLA-ave(ks) in the inductance LA andthe condenser C due to iLA-ave(ks) as the average detected current inthe control cycle (ks), as well as a ratio for the input voltage Vin ofeach voltage component of the voltage component ((Ts+Td)²/2C)·iR(ks) inthe condenser C due to an input voltage (Ts+Td)·vo(ks), and a term(Td/Ts)·ΔT(k−1) of the pulse width ΔT(k−1) in previous cycle (k−1).

In regard of the inductance, capacitance and the ratio for the inputvoltage Vin of each of a voltage component of each element of aresistance, the pulse width ΔT(k) has a term of time (Ts+Td) or (Ts+Td)²obtained by adding the delay time Td to the control cycle Ts.Furthermore, the previous pulse width ΔT(k−1) has a coefficient of theterm of (Td/Ts) obtained by dividing the delay time Td by the controlcycle Ts.

The pulse width ΔT(k) in Formula 5 shows the case where the averagevalue of the inductance current is used as the detection value. InFormula 5, iLA-ave(ks) is the average value of the inductance currentiLA and Vo(ks) is the detected output voltage vo. The average currentiLA-ave of the inductance current at the point ks uses an average valueof a period of [ks−1−ks], which is one cycle of the switching, and anaverage value input of [k−k+1] derived at the point ks uses the valueVin·Duty(=ΔT(k)/(Ts).

The formula for the discrete control using the detection value at eachpoint in the control cycle can be obtained in Formula 5 by using thedetection value at each point in the control cycle instead of theaverage value.

Formula 5 is an approximate expression in which the general solutionx(t) of the state equation is approximated by a quadratic expansionexpression, and includes a quadratic term for (Ts+Td) that includes thedelay time Td. The approximation of the general solution x(t) of thestate equation by a higher-order expansion expression can increase thedegree of approximation of the width ΔT(k).

The pulse width ΔT(k) expressed by Formula 5 includes the term of thedelay time Td. By opening and closing the switching device of the maincircuit based on the pulse width ΔT(k) including the term of the delaytime Td, the control considering the delay time Td is implemented.

<Delay>

A description will be made on the delay time Td in the control cycle ksof the control unit (controller) with respect to the cycle k of the maincircuit by referring to FIGS. 4 to 7 .

<When there is No Delay>

FIG. 4 shows the case where no delay time Td occurs. In which case, thecycle k of the main circuit matches the control cycle ks of the controlunit (controller).

FIGS. 4(a) to 4(d) respectively show the sampling, the detected output,the command value and the manipulated variable in the control unit(controller), and FIGS. 4(e) and 4(f) respectively show the gate signaland the output in the main circuit.

Since the cycle k of the main circuit matches the control cycle ks ofthe control unit (controller), no time lags occur between the detectedoutput (b) and the output (f). At the point ks, the pulse width ΔT(k) asthe manipulated variable is formed on the basis of the detected output(b) and the command value (c). The pulse width ΔT(k) is the manipulatedvariable for controlling the output at the point (k+1) to reach thecommand value. The main circuit performs the closing operation of theswitching device with the pulse width ΔT(k) at the point ks in thecontrol cycle between the point (k) and the point (k+1), therebycontrolling the output to reach the command value at the point (k+1).

<When there is a Delay (but not Considering the Delay Time Td)>

FIG. 5 shows the case where there is a delay, but the delay time Td isnot considered for the pulse width ΔT(k) as the manipulated variable inthe discrete control. The delay causes a shift for the delay time Tdbetween the cycle k of the main circuit and the control cycle ks of thecontrol unit (controller).

FIGS. 5(a) to 5(d) respectively show the sampling, the detected output,the command value and the manipulated variable in the control unit(controller), and FIGS. 5(e) and 5(f) respectively show the gate signaland the output in the main circuit.

There is a shift of the delay time Td between the cycle k of the maincircuit and the control cycle ks of the control unit (controller), andthe detected output (b) of the control unit (controller) is detectedwith a delay of the value Td from the output (f) of the main circuit.The illustrated detected output (b) shows, for the point ks when thereis no delay, an output waveform at the point ks which is previous pointby the delay time Td.

The pulse width ΔT(k) thus formed is a manipulated variable that makesthe output to follow the command value at the point ks based on thedetected output (b) and the command value (c) at the point ks. In theformation of the pulse width ΔT(k), the control cycle ks of the controlunit (controller) lags behind the cycle k of the main circuit by thedelay time Td, resulting in the occurrence of a detection error as shownin the detected output (b) between the output and the detected output. Adetection error shown in FIG. 5(b) is a difference between the output,depicted by a square in the figure, and the detected output, depicted bya cross in the figure, and thus represents a detection error at thepoint (ks+1).

The main circuit obtains at the point k the pulse width ΔT(k) obtainedat the point ks, which lags by the delay time Td, in the control cyclebetween the point k and the point (k+1), so as to perform the closingoperation of the switching device. Since the pulse width ΔT(k) as themanipulated variable of the main circuit is formed based on the detectedoutput including an output error caused by the delay time Td, sufficienttime cannot gained to complete the gate control of the pulse width ΔT(k)between the points [ks−ks+1], and consequently the output controlled bythe pulse width ΔT(k) has an output error (FIG. 5(f)) with respect tothe command value. The error in the detection value and the error in theoutput become factors for oscillation.

<When there is a Delay (Considering the Delay Time Td)>

FIG. 6 shows the case where there is a delay and the delay time Td isconsidered for the pulse width ΔT(k) as the manipulated variable in thediscrete control. An acquisition delay occurring in the detector whenobtaining the detection values of the voltage and the current, acomputation delay occurring during the computation of the manipulatedvariables for the discrete control and others, and a delay, such asresponse delay, in the switching device of the DC/DC converter cause ashift in the delay time Td between the cycle k of the main circuit andthe control cycle ks of the control unit (controller).

As with the case of the control with no delay shown in FIG. 4 , if thepoint (ks+1) is predicted by using the value at the point ks, an erroroccurs in the manipulated variable because the detected output forobtaining the manipulated variable includes a delay amount due to theshift in the delay time Td.

Thus, in order to prevent the error due to the delay time Td, thecontrol according to the present invention makes a prediction in thecontrol cycle period [ks−ks+1] of the control unit about the point((ks+1)+Td) after the delay time Td from the point (ks+1), instead ofthe point (ks+1) using the value of the detection signal at the pointks. This predicted point is the point (k+1) in the control cycle of themain circuit. The discrete control at the point ks obtains themanipulated variable (pulse width ΔT(k)) for allowing an object to becontrolled to follow the command value at the point (k+1) which is aftera lapse of the time ((ks+1)+Td) from the point ks.

FIGS. 6(a) to 6(d) respectively show the sampling, the detected output,an estimated output (average value), the command value and themanipulated variable of the control unit (controller), and FIGS. 6(e)and 6(f) respectively show the gate signal and the output in the maincircuit.

Since there is a lag of the time delay Td between the cycle k of themain circuit and the control cycle ks of the control unit (controller),the detected output (b) is detected with a delay of Td from the output.The detected output in FIG. 6(b) (shown by a thick line) indicates thestate where the detected output is detected with the lag of the delaytime Td behind the output (shown by a thin line).

The present invention forms the pulse width ΔT(k) as a manipulatedvariable at the point (k+1) based on the detected output (b) at thepoint ks and the command value (c), as well as the previous pulse widthΔT(k−1).

Since the detection value at the point ks used by forming the pulsewidth ΔT(k) at the point k is a value with the lag of the delay time Td,not the value at the point k, the delay time Td is compensated withrespect to the previous pulse width ΔT(k−1) and the detection value atthe point ks so as to obtain the value at the point k that is a pointafter the delay time Td from the point ks. The compensation of the delaytime Td in the pulse width ΔT(k) can eliminate the detection errorcaused by the delay time Td.

Due to the delay time Td occurring in the control unit, the pulse cycleks on the control side is recognized by the main circuit side asdelaying by the time Td, while the cycle k on the main circuit side isrecognized by the control side as advancing by the delay time Td.

In this way, since the detection value at the point ks of the controlunit delays by the delay time Td as viewed from the main circuit side,the detection value of the control unit has the delay by the delay timeTd at the point k of the main circuit. The present invention obtains thevalue by compensating the delay time Td with respect to the previouspulse width ΔT(k−1) and the detection value at the point ks on thecontrol side, thereby using the compensated value to form the pulsewidth ΔT(k) at the point k. The pulse width ΔT(k) is a manipulatedvariable for performing control in the cycle k from the point k of themain circuit.

Although the main circuit advances by the delay time Td viewed from thecontrol side, the error caused by the delay time Td is eliminated bycontrolling the gate using the manipulated variable of the pulse widthΔT(k) in which the delay time Td is compensated. In this case, the cyclek of the main circuit is a period between the point k and the point(k+1), and the gate is controlled by the manipulated variable of thepulse width ΔT(k) in this period.

In this context, the detection value to be used for forming the pulsewidth ΔT(k) between [k−k+1] in the cycle k applies the detection valueat point ks which is before the point k by the delay time Td as thevalue at the point k without being processed, as well as an estimatedvalue obtained from a value within a predetermined period in the pulsecycle ks before the point ks. The use of the estimated value obtainedfrom the value within the predetermined period enables to avoid thedetection error caused by the fluctuation in the detection value withinthe pulse cycle ks.

In the estimation of the detection value using the value within thepredetermined period, for instance, an average value of an detectionvalue of the capacitance current ic in the period of [ks−1−ks] iscomputed, and then the estimated value according to the average value isdefined as the detection value of the capacitance current ic at thepoint ks, so as to use the detection value to form the pulse width ΔT(k)for performing the gate control in the period of [k−k+1] in the cycle k.

In the formation of the pulse with ΔT(k), although the control cycle ksof the control unit (controller) lags behind the cycle k of the maincircuit by the delay time Td, the prediction is made on the point (k+1)in the control cycle of the main circuit which is the point ((ks+1)+Td)behind the point (ks+1) by the delay time Td, not on the point (ks+1).The main circuit performs the closing operation of the switching deviceby the pulse width ΔT(k) obtained at the point ks which lags by thedelay time Td in the control cycle [k−k+1] between the point k and thepoint (k+1).

The pulse width ΔT(k) expressed by the above Formula 5, which is themanipulated variable of the discrete control considering the delay timeTd, takes account of the term of time (Ts+Td) of the control cycle Tsthat is equivalent to the delay time Td and the point (k+1) in thecontrol cycle.

On the control unit (controller) side, a value detected at a point,which lags behind the point ks by the delay time Td, matches the cycle kon the main circuit side. Thus, when viewed from the main circuit side,the manipulated variable obtained in the cycle k of the main circuit(pulse width ΔT(k)) is a manipulated variable that recognizes accuratelythe state of the main circuit, and consequently the error between themain circuit and the control side due to the delay time Td iseliminated.

This allows enough time for the gate control of the pulse width ΔT(k),thereby suppressing the output error between the output controlled bythe pulse width ΔT(k) and the command value (circle drawn by a dashedline in FIG. 6 ). FIG. 6(b) shows a detection error (difference betweenthe output and the detected output) at the point (ks+1).

(When delay time Td exceeds one control cycle Ts) The derivation of theabove expression of the discrete control considering the delay time Tdhas been described on the assumption that the delay time Td is withinone sampling cycle Ts in the cycle k of the main circuit. Alternatively,the manipulated variable (pulse width ΔT(k)) can be determined by asimilar technique by extending the predicted point according to thedelay time Td when the delay time Td exceeds one sample cycle Ts of thecycle k of the main circuit.

FIG. 7(a) illustrates a case where the delay time Td is within onesampling cycle Ts, and FIG. 7(b) illustrates a case where the delay timeTd is over one sampling cycle Ts and within two sampling cycle 2 Ts.

As shown in FIG. 7(a), when the delay is within one sampling cycle Ts inthe control cycle ks, a prediction is made based on the detected outputat the point ks and the command value about the point (k+1) in thecontrol cycle of the main circuit, which is the point (Ts+Td) after thepoint ks by the delay time Td in one sampling cycle Ts. In the case ofthe multi-phase interleaving, a prediction is also made about the point(k+1) in the control cycle of the main circuit, which is the point(Ts+Td) after the point ks by the delay time Td in one sampling cycleTs.

The duration [ks−k+1] is used as an integration duration in the stateequation with respect to the point ks to predict the point (k+1) after(Ts+Td), in order to perform the switching control with the pulse widthΔT(k) in a control duration of [k−k+1].

In FIG. 7(b), when the delay is over one sampling cycle Ts and withintwo sampling cycles 2T sin the control cycle ks, a prediction is madebased on the detected output at the point ks and the command value abouta point (k+2) in the control cycle of the main circuit after the pointks by (Ts+Td) obtained by adding the delay time Td and the samplingcycle Ts. Since the delay time Td exceeds one sampling cycle Ts, thepoint after the point ks by (Ts+Td) becomes the point (k+2).

The duration [ks−k+2] is used as an integration duration in the stateequation with respect to the point ks to predict the point (k+2) after(2 Ts+Td), in order to perform the switching control with the pulsewidth ΔT(k) in a control duration of [k+1−k+2].

(Three-Phase Interleaving Discrete Control)

In the single-phase discrete control described above, the manipulatedvariable for the single-phase discrete control has been presented. Now,in the case of employing a multi-phase interleaving system that is atechnique for increasing the speed of a DC/DC converter in order tospeed-up the DC/DC converter, a description will be made about athree-phase discrete control that enhances a manipulated variable forthe discrete control of a three-phase interleaved step-down DC/DCconverter.

FIG. 8 illustrates the application of the three-phase interleavingsystem as multi-phase interleaving system, showing an example of a pulsewidth ΔT(k) for a three-phase current.

In the three-phase interleaving system, each phase of the three-phase isshifted by 120 degrees to triple a ripple frequency. Thus, in thethree-phase interleaving system, an output ripple equivalent to that inthe single-phase system can be achieved with one-third of the outputcapacitor's ability, thereby increasing the speed of an operation ofswitching to the voltage level of the DC/DC converter.

FIG. 8(a) illustrates that the pulse widths ΔT(k) of three phasecurrents in the three-phase current are overlapped one another in thetime width T in one cycle of the switching. FIG. 8(b) illustrates thatthe pulse widths ΔT(k) of two phases currents in the three-phase currentare overlapped each other in the time width Ts in one cycle of theswitching. FIG. 8(c) illustrates that there is no overlapping of thepulse widths ΔT(k) of the phase currents regarding the three-phasecurrent.

When the switching operation is carried out on the switching circuit 3by the multi-phase interleaving of n-phase, the n-inductances L (L1 toLn) included in the LC chopper circuit of the main circuit 2 are fedwith inductance currents iL1 to iLn, respectively. The control unit 6inputs as control current a current containing a combined current iLwhich is obtained by combining the phase current values of theinductance currents iL1 to iLn.

The control current may apply a capacitance current iC, obtained bysubtracting the load current iR from the combined current iL, inaddition to the combined current iL obtained by combining the inductancecurrents of the phase currents.

In the single-phase discrete control, the manipulated variable for thediscrete control by taking into consideration the delay time Td isexpressed by Formula 5. By extending the pulse width ΔT(k) to thethree-phase interleaved step-down DC/DC converter, the manipulatedvariable for the discrete control considering the delay time Td can beobtained in a converter with the increasing speed by the three-phaseinterleaving system. Here, the description refers to the three-phaseinterleaving system by way of example of the multi-interleaving, andthus the pulse width can also be applied to other multi-interleavingsystem with more than three phases.

FIG. 9 shows a schematic configuration of the step-down DC/DC converterby the three-phase interleaving system. Switching devices S1A, S2A andan inductance LA, switching devices S1B, S2B and an inductance LB, aswell as switching devices S1C, S2C and an inductance LC respectivelyconfigure each phase of the three phases, and have a common capacitanceC and load resistance RL.

Expressions of the constant-current control for detecting the combinedcurrent as control current and of the control current and the outputvoltage for the constant-voltage control are derived. FIGS. 10(a) and10(b) show equivalent circuits for the three-phase interleavingstep-down DC/DC converter circuit shown in FIG. 9 , which representequivalent circuits for time-bandwidth that is sufficiently longer thana switching frequency in the region of a closed-loop automaticcontrolled response.

<Constant-Voltage Control>

The equivalent circuit of an LCR circuit in FIG. 10(b) illustrates theconstant-voltage control for detecting a detected output voltage vo. Inthis figure, shown is an example of a DC/DC converter that includes astep-down chopper circuit consisting of the LCR circuit.

In the equivalent circuit of the LCR circuit, the detected outputvoltage vo obtained in a step response by inputting an input voltage Uis expressed by the following formula.

$\begin{matrix}\left( {{Formula}15} \right) & \end{matrix}$ $\begin{matrix}{{v_{o}(s)} = {{\frac{\frac{R}{1 + {sCR}_{L}}}{{{sL}/3} + \frac{R_{L}}{1 + {sCR}_{L}}}\frac{U}{s}} = {{\frac{\frac{3}{LC}}{s^{2} + {\frac{1}{{CR}_{L}}s} + \frac{3}{LC}}\frac{U}{s}} = {\frac{\omega_{n}^{2}}{s^{2} + {2{ϛ\omega}_{n}s} + \omega_{n}^{2}}\frac{U}{s}}}}} & (6)\end{matrix}$

The above Formula 6 indicates that the detected output voltage vo is asecondary oscillation voltage, which suggests the occurrence of anovershoot and an undershoot.

<Constant-Current Control>

In the equivalent circuit in FIG. 10(a), a combined current(iLA+iLB+iLC=iL) composed of the phase currents iLA, iLB and iLC of eachphase is represented as a current source, and a combined inductanceconsisting of the inductances L of three switching circuits isrepresented as (L/3). In this equivalent circuit, the step response ofthe detected output voltage vo due to the input current (iL) from thecurrent source is expressed by the following formula.

$\begin{matrix}\left( {{Formula}16} \right) & \end{matrix}$ $\begin{matrix}{{{v_{o}(s)} = {{\frac{R_{L}}{1 + {sCR}_{L}}\frac{i_{L}}{s}} = {{Ri}_{L}\left( {\frac{1}{s} - \frac{1}{s + {1/{CR}_{L}}}} \right)}}}{{v_{o}(t)} = {R_{L}{i_{L}\left( {1 - e^{{- \frac{1}{{CR}_{L}}}t}} \right)}}}} & (7)\end{matrix}$

Formula 7 indicates that the step response of the detected outputvoltage vo increases in an exponential manner toward (RL·iL) withoutproducing the secondary oscillation voltage.

A time function iL(t) of the combined current of the inductance currentiL is defined by the following Formula 8.{Formula 17}i _(L)(t)=i _(C)(t)+i _(R)(t)=A _(V) {V _(REF) −v _(o)(t)}+i_(R)(t)  (8)

A combined current (iL(t)), a capacitance current iC(t) and a detectedoutput voltage vo(t) are expressed by the following Formula 9.

$\begin{matrix}\left( {{Formula}18} \right) & \end{matrix}$ $\begin{matrix}{{{i_{L}(t)} = {{{i_{C}(t)} + {i_{R}(t)}} = {{A_{V}\left\{ {V_{REF} - {v_{o}(t)}} \right\}} + {i_{R}(t)}}}}{{i_{C}(t)} = {{A_{V}\left\{ {V_{REF} - {v_{o}(t)}} \right\}} = {C\frac{d}{dt}{v_{o}(t)}}}}{{v_{o}(t)} = {V_{REF}\left\{ {1 - e^{{- \frac{A_{v}}{C}}t}} \right\}}}} & (9)\end{matrix}$

The detected output voltage vo(t) expressed by Formula 9 indicates thatthe load resistance RL is removed from the detected output voltage vo(t)expressed by Formula 7, and a final value after a lapse of sufficienttime (t→∞) converges at a command voltage Vref.

Thus, the constant-current control using the combined current of theinductance current iL(t) shown in Formula 8 as control current enablesto control the step response without producing the secondary oscillationvoltage.

In the detected output voltage vo(t) expressed by Formula 9, Av is acoefficient with which a difference value (Vref−Vo(t)) between thedetected output voltage vo(t) and the command voltage Vref ismultiplied. For example, the larger the coefficient Av, more stronglythe step response reflects the difference value (Vref−Vo(t)).

<State Equation of Bilateral Step-Down Chopper Circuit>

Next, a state equation of a bilateral step-down chopper circuit by thethree-phase interleaving system is derived. FIG. 11 shows an equivalentcircuit of one of three phases. In order to convert the combined current(iL) expressed by the above Formal 8 into a form suitable for theconstant-current control, the state equation of the combined currentiL(=iL1+iL2+iL3) obtained from values iL1, iL2 and iL3 shown in FIG. 9is determined to derive a relational expression with respect to thepulse width ΔT.

The ON/OFF operations are performed on the phases S1A to S1C and S2A toS2C shown in FIG. 9 , so that a voltage of Vin or zero is applied acrossu1(τ), u2(τ) and u3(τ). If the superposition theory is applied, u1(τ) isrepresented by an equivalent circuit shown in FIG. 11 . In FIG. 11 ,when the device S1A is turned on and the device S2A is turned off, u1(τ)becomes Vin, and when the device S1A is turned off and the device S2A isturned on, u1(τ) becomes zero. In this regard, the inputs Vin in thedevices S1B and S2B as well as the devices S1C and S2C are in a shortcircuit condition.

Provided that LA=LB=LC=L, the state equations for the voltages u1(t),u2(t) and u3(t) are determined in FIG. 11 , so as to obtain the stateequation for the three-phase interleaved step-down DC/DC converteraccording to the theory of the superposition of these voltages.(Formula 19){dot over (x)}(t)=A ₂ x(t)+B ₂ u(t)  (10)

In this regard, x(t) represents currents iLA(t), iLB(t) and iLC(t) ofthe inductances LA, LB and LC, respectively, and is also an element ofthe detected output voltage vo(t), u(t) represents input voltages u1(t),u2(t) and u3(t) of the respective phases, A2 represents the term of acoefficient consisting of the elements of the inductance L, thecapacitance C and the resistance R in each phase, and B2 represents theterm of a coefficient consisting of the element of the inductance L ineach phase.

As with the case of the above-described single-phase, provide that aduty in the period of [(ks−1)−k] is ΔT(k−1)/Ts and a duty in the periodof [k−(k+1)] is ΔT(k)/Ts, the pulse width ΔT(k) as manipulated variableis computed by the following Formula 11.

$\begin{matrix}\left( {{Formula}20} \right) & \end{matrix}$ $\begin{matrix}{{{\Delta T}(k)} = {\frac{{\frac{L}{3}{i_{L}\left( {k + 1} \right)}} - {\left\{ {\frac{L}{3} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{L - {ave}}\left( k_{s} \right)}}}{V_{in}} + \frac{{\left( {T_{s} + T_{d}} \right){v_{o}\left( k_{s} \right)}} - {\frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}{i_{R}\left( k_{s} \right)}}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}} & (11)\end{matrix}$

Thus, the expression of the discrete control considering the delay timeTd based on the manipulated variable as ΔT(k) and the command value asiL(k+1) is derived by Formula 5 in the case of the single-phase andFormula 11 in the case of the three-phase.

The pulse width ΔT(k) of n-phase is expressed by the following formula.

$\begin{matrix}\left( {{Formula}21} \right) & \end{matrix}$ $\begin{matrix}{{{\Delta T}(k)} = {\frac{{\frac{L}{n}{i_{L}\left( {k + 1} \right)}} - {\left\{ {\frac{L}{n} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{L - {ave}}\left( k_{s} \right)}}}{V_{in}} + \frac{{\left( {T_{s} + T_{d}} \right){v_{o}\left( k_{s} \right)}} - {\frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}{i_{R}\left( k_{s} \right)}}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}} & (12)\end{matrix}$

(In the Case of Multi-Phase Interleaving)

In the above description, the switching operation is conducted by usingthe single-phase gate signal. Correspondingly, the switching operationby the multi-phase interleaving can be performed with a multi-phase gatesignal.

As with the switching operation by the single-phase gate signal, whenthe delay time Td is within the sampling cycle Ts, the multi-phaseswitching operation makes a prediction based on the detected output andthe command value at the point ks about the point (k+1) in the controlcycle of the main circuit, which is the point of (Ts+Td) from the pointks obtained by adding the delay time Td to the sampling cycle Ts.

In the multi-phase switching operation, if an estimated value accordingto the average value is used as the detection value at the point ks tobe used for computation in the computation of the pulse width ΔT(k) asthe manipulated variable, a ripple cycle Tr is adopted in an averageperiod, instead of the sampling cycle Ts. For example, in thethree-phase interleaving consisting of A-phase, B-phase and C-phase, ifthe delay time Td is shorter than the sampling cycle Ts, a prediction ismade at a point ksA in the A-phase about a point kA+1 of the maincircuit by taking account of the delay by the delay time Td to therebydetermine the pulse width ΔT(k) in the period [kA−kA+1] in the cycle kof the main circuit cycle.

In this case, an average value in a period of a control cycle[ksC−1−ksA] is used as the detection value at the point ksA in thecondenser current iC. In this context, the point ksC−1 is a point in theC-phase in the cycle ks, and time width of the control cycle is oneripple cycle Tr(=Ts/3). In the n-phase, one ripple cycle Tr is Ts/n.

At the sampling point ks, the delay time Td is considered to predict thepoint (k+1) in the main circuit so as to determine the pulse width ΔT(k)of [k−k+1]. At this time, if the average value is applied as detectionvalue, an average value in a period before the point k by (ripple cycleTrof the single-phase+delay time Td).

Moreover, when the delay time Td is one sampling cycle Ts or more andtwo sampling cycles 2 Ts or less, a point (k+2) in the control cycle ofthe main circuit that is a point (nTs+Td) after the time delay Td in are-sampling cycle Ts from the point ks is predicted based on thedetected output or estimated value at the point ks and the commandvalue. In this case, since the delay time Td exceeds one sampling cycleTs, the point after (nTs+Td) from the point ks becomes the point (k+2).

In the case of the single phase, the pulse width ΔT(k) is determined forevery sampling cycle Ts. In the case of the three phases, if the timedelay Td is in [0−Ts], the pulse width ΔT(k) of [k−k+1] is determined atthe point ks in the same way as the case of the single phase, and if thedelay time Td is in [Ts−2 Ts], the pulse width ΔT(k) of [k+1−k+2] isdetermined at the point ks.

In the discrete control by the three-phase interleaving system, as withthe case of the single-phase discrete control, the detection value ofthe inductance current iLA(t) is obtained from the average value inorder to suppress the influence by the fluctuation in the detectionvalue due to the ripple component contained in the inductance current tobe detected.

A period during which the current fluctuates periodically corresponds toone cycle of the ripple frequency. This period of one cycle of theripple frequency is defined as an acquisition period for acquiring theaverage value, so that the fluctuation caused by the ripple component isaveraged. In the case of the single phase, the acquisition period foracquiring the average value of the detection value is defined to be onecycle of the cycle k of the main circuit which is one cycle of theripple frequency. One cycle of the cycle k of the main circuit is onecycle of the switching operation of the switching device in the maincircuit.

In the three-phase interleaving system, the above-described period isalso defined to be one cycle of the ripple frequency. One cycle of theripple frequency is ⅓ of the cycle k of the main circuit for theswitching in the case of the three-phase system. Correspondingly, in themulti-phase interleaving of n-phase, which is more than three phases,the average value is determined by considering the acquisition period as1/n cycle of the cycle k of the main circuit for switching.

FIG. 12(a) shows an ON/OFF state of a three-phase switching circuit,FIG. 12(b) shows inductance currents iL1 to iL3 flown by thesingle-phase switching, and FIG. 12(c) shows the combined current(iL1+iL2+iL3) flown by the three-phase switching. A period for obtainingan average current is one cycle of the switching cycle in thesingle-phase, whereas the period in the three-phase is ⅓ of theswitching cycle.

(Discrete Control by Capacitance Current iC)

Formula 5 for the single-phase discrete control and Formula 12 for themulti-phase discrete control by considering the delay time Td derive theinductance current iL(k+1) as command value.

However, there are possible delay times beyond the controlling cycle ina DC current sensor for detecting the inductance current iL and anisolated amplifier for detecting the detected output voltage vo. In sucha case, it is not expected that the pulse width ΔT(k) derived by Formula5 or Formula 12 will attain the stability adequate to the switchingoperation.

(Discrete Control Using Capacitor Current iC as Detection Value)

In order to eliminate an excessive delay caused by using the inductancecurrent iL as detection value, the inductance current iL detected by theDC current sensor is replaced with the capacitance current iC detectedby an AC current sensor to conduct the discrete control on a controlsystem using the capacitance current iC as the detection value. As theAC current sensor can perform fast detection, the occurrence of delay isreduced in the detection.

Now, a description will be made on the combination of theconstant-current control and the constant-voltage control in thediscrete control of the switching operation of the voltage level of theDC/DC converter.

First, a description will be made about the constant-voltage discretecontrol, the constant-current discrete control, and the combination ofthe constant-voltage discrete control and the constant-current discretecontrol in the case of where no delay occurs, and then a descriptionwill be made about a modal control of the discrete control in the caseof where a delay occurs. In regard of the modal control of the discretecontrol, a description will be made on the control by taking intoconsideration the overshoot and the undershoot occurring duringswitching from the constant-current control to the constant-voltagecontrol as well as the delay time. The following description about eachtype of discrete control illustrates High/Law two-level control thatswitches the voltage level between a high voltage level (High level) anda low voltage level (Low level).

FIGS. 12(b) and 12(c) schematically show current waveforms forillustrative purposes, and do not show actual waveforms.

<Controlling Form by Combination of Constant-Voltage Control andConstant-Current Control>

A description will be made about a controlling form by the combinationof the constant-voltage control and the constant-current control in theHigh/Low two-level control, by referring to FIG. 13 .

In this controlling form, the constant-voltage control and theconstant-current control are carried out in combination to effect atransition between two levels for switching a power level from a lowpower side to a high power side and switching the power level from thehigh power side to the low power side.

FIG. 13 illustrates the controlling form by the combination of theconstant-voltage control and the constant-current control, wherein FIG.13(a) shows a schematic configuration of the control unit, FIGS. 13(b)and 13(c) show the command voltage Vref and a command current IC-ref,and FIG. 13(d) shows the detected output voltage vo. In this figure, thecapacitance current iC is used as the detected current.

The constant-voltage control is performed in the retention period inwhich the command voltage is retained on the low power side and the highpower side, and the constant-current control is performed in thetransition period in which the power level is switched from the lowpower side to the high power side.

The constant-voltage control uses the pulse width ΔT(k) expressed by thefollowing Formula 13 or 14 to retain the command voltage Vref.

$\begin{matrix}\left( {{Formula}22} \right) & \end{matrix}$ $\begin{matrix}{{{\Delta T}(k)} = {\frac{1}{V_{in}(k)}{\frac{L}{3}\left\lbrack {{A_{v}V_{ref}} - {\left( {1 - \frac{3{Ts}^{2}}{2{LC}}} \right){i_{c}(k)}} + {\left( {{\frac{3}{L}T_{s}} - A_{v}} \right){V_{o}(k)}}} \right\rbrack}}} & (13)\end{matrix}$ $\begin{matrix}\left( {{Formula}23} \right) & \end{matrix}$ $\begin{matrix}{{{\Delta T}(k)} = {\frac{V_{ref} - {\left\{ {\frac{L}{3T_{s}} - \frac{T_{s}}{2C}} \right\}{i_{c}(k)}}}{V_{in}(k)}T_{s}}} & (14)\end{matrix}$

The pulse width ΔT(k) expressed by Formula 13 uses the detectedcapacitance current iC(k) and the detected output voltage vo(k) toconduct the control in such a way that the detected output voltage vo(k)becomes the command voltage Vref.

The pulse width ΔT(k) expressed by Formula 14 uses the detectedcapacitance current iC(k) to conduct the control in such a way that thedetected output voltage vo(k) becomes the command voltage Vref. InFormula 14, the coefficient Av is set to Av=3 Ts/L to eliminate the needfor detecting the detected output voltage vo(k), and thus only thecapacitance current iC(k) is detected to determine the pulse widthΔT(k).

In the constant-current control, the pulse width ΔT(k) expressed by thefollowing Formula 15 is used to keep the detected current iC as thecommand current IC-ref so as to shift from the command voltage VL on thelow power side toward the command voltage VH on the high power side orfrom the command voltage VH on the high power side toward the commandvoltage VL on the low power side.

$\begin{matrix}\left( {{Formula}24} \right) & \end{matrix}$ $\begin{matrix}{{{\Delta T}(k)} = \frac{{\frac{L}{3}I_{c - {ref}}} - {\left\{ {\frac{L}{3} - \frac{T_{s}^{2}}{2C}} \right\}{i_{c}(k)}} + {T_{s}{v_{o}(k)}}}{V_{in}(k)}} & (15)\end{matrix}$

As to the command current IC-ref of the capacitance current in theHigh/Low two-level control, the command current of IC-refH thatcorresponds to VH of the High level and the command current of IC-refLthat corresponds to VL of the Low level are taken as examples.

In the High/Low two-level control, the constant-voltage control in thepower retention period and the constant-current control in the powertransition period are repeated.

(Mode Controls in Discrete Control)

In the discrete control, when the control form by the combination of theaforementioned constant-voltage control and the constant-current controlis adopted, it is necessary to take into consideration the delay time aswell as the overshoot and undershoot occurring during switching from theconstant-current control to the constant-voltage control. Now, adescription will be made about mode controls in the discrete controlconsidering the delay time and the overshoot and undershoot.

The High/Low pulse operation performs the discrete control by aplurality of modes with the combinations of the constant-current controland the constant-voltage control, in order to take place of the smoothtransition between the high power side and the low power side. In theswitching operation during control switching from the constant-currentcontrol to the constant-voltage control in the control with thecombination of the above-described constant-voltage control and theconstant-current control, the discrete control according to the presentinvention utilizes three modes for controlling as described below so asto conduct the switching operation smoothly by suppressing the overshootand undershoot.

In the High/Low pulse operation, the discrete control of the presentinvention consists of, as shown in FIG. 14 , a first mode (mode I) forthe constant-current control to be applied in the voltage transitionbetween the high power side and the low power side, a third mode (modeIII) for the constant-voltage control to be applied during the operationwith the high power or low power, as well as a second mode (mode II) forthe control in a buffer period for smoothly switching from theconstant-current control to the constant-voltage control. In thefollowing description, the first mode, second mode and third mode arerepresented as the mode I, mode II and mode III, respectively.

FIG. 14 shows a state of power transition by three modes of the mode Ito mode III in the discrete control of the present invention, in whichthe respective modes in the High/Low pulse operation are illustrated.FIG. 14(a) shows the power transition from the low power side to thehigh power side, and FIG. 14(b) shows the power transition from the highpower side to the low power side.

When performing the transition of the power from the low power side tothe high power side, the constant-voltage control in the mode III iscarried out during the low power operation, and the constant-currentcontrol in the mode I is carried out during the voltage transition fromthe low power side to the high power side, and the constant-voltagecontrol of the buffer mode in the mode II is carried out between theswitching from the constant-current control in the mode I to theconstant-voltage control in the mode III.

On the other hand, when performing the transition of the power from thehigh power side to the low power side, the constant-voltage control inthe mode III is carried out during the high power operation, and theconstant-current control in the mode I is carried out during the voltagetransition from the high power side to the low power side, and theconstant-voltage control of the buffer mode in the mode II is carriedout between the shift from the constant-current control in the mode I tothe constant-voltage control in the mode III.

Next, the mode I, mode II and mode III will be described.

<Mode I: Constant-Current Control>

The mode I is the constant-current discrete control to be conductedduring the voltage transition between the high power side and the lowpower side, which control is applied to the transition from low powerside to the high power side and the transition from the high power sideto the low power side. The constant-current discrete control isimplemented to prevent the occurrence of overshoot and undershoot andthe occurrence of overcurrent during the transition.

When the inductance current iL(ks) is detected by the DC current sensor,the delay time of several μs occurs. By contrast, concerning the ACcurrent sensor, many general-purpose devices have less delay time. Thus,in order to use the capacitance current that can be detected by the ACcurrent sensor for control, the command value iL(k+1) related to theinductance current iL is defined by the following Formula 16. In thefollowing description, the command value of the voltage is presented asVref, and the command value of the capacitance current is presented asIC-ref or ICref.(Formula 25)i _(L)(k+1)=I _(Cref) +i _(R)(k _(s))  (16)

Furthermore, the relation between an average current iL-ave(ks) of theinductance, an average current iC-ave(ks) of the capacitance current iCand the load current iR(ks) is expressed by the following Formula 17.(Formula 26)i _(L-ave)(k _(s))=I _(C-ave)(k _(s))+i _(R)(k _(s))  (17)

By substituting Formula 16 and Formula 17 into Formula 13, theexpression of the discrete control using the detected capacitancecurrent iC can be obtained.

$\begin{matrix}\left( {{Formula}27} \right) & \end{matrix}$ $\begin{matrix}{{{\Delta T}(k)} = {\frac{{\frac{L}{3}I_{Cref}} - {\left\{ {\frac{L}{3} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{C - {ave}}\left( k_{s} \right)}}}{V_{in}} + \frac{\left( {T_{s} + T_{d}} \right){v_{o}\left( k_{s} \right)}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}} & (18)\end{matrix}$

The pulse widths ΔT(k) in the single-phase and n-phase controls areexpressed by the following formulas.

$\begin{matrix}\left( {{Formula}28} \right) & \end{matrix}$ $\begin{matrix}{{{\Delta T}(k)} = {\frac{{LI}_{Cref} - {\left\{ {L - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{C - {ave}}\left( k_{s} \right)}}}{V_{in}} + \frac{\left( {T_{s} + T_{d}} \right){v_{o}\left( k_{s} \right)}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}} & (19)\end{matrix}$ $\begin{matrix}\left( {{Formula}29} \right) & \end{matrix}$ $\begin{matrix}{{{\Delta T}(k)} = {\frac{{\frac{L}{n}I_{Cref}} - {\left\{ {\frac{L}{n} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{C - {ave}}\left( k_{s} \right)}}}{V_{in}} + \frac{\left( {T_{s} + T_{d}} \right){v_{o}\left( k_{s} \right)}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}} & (20)\end{matrix}$

The pulse widths ΔT(k) obtained by Formula 19 and Formula 20 include asthe detection values the capacitance average current iC-ave(ks) and thedetected output voltage vo(ks).

The calculation of the capacitance average current iC-ave(ks) isimplemented in such a way that the average value is determined bydetecting the capacitance current iC for every sampling cycle Tsamplewhich is shorter than the control cycle Ts to thereby obtain the averagecurrent in the control cycle Ts.

Since the detected voltage vo(ks) of the output is typically obtainedvia an insulated amplifier, the delay time of several μs occurs. Thus,the detected output voltage vo(ks) is determined by using thecapacitance current detected by the AC current sensor that is capable ofhigh-speed detection.

The calculation of the detected output voltage vo(ks) using thecapacitance current iC is implemented in such a way that the detectionand the computation are performed for every sampling cycle Tsample whichis shorter than the sampling cycle Ts to obtain the average current inthe sampling cycle Ts and the change in voltage using the averagecurrent.

Provided that the detection value of the output voltage obtained at thetime immediately before the shift either from the low power side to thehigh power side or from the high power side to the low power side isdefined to an initial value vo(ks), an output voltage vodet(ks+1) can beobtained by the computation in the control sample after the samplingcycle Ts.

The output voltage vodet(ks+1) is represented by the sum of valuesobtained by multiplying each of a voltage i_(c)/C in the sampling cycleTs of the capacitance C and the output voltage vodet(ks) at the point ksby a coefficient of (ks+(m−1)·Tsample/Ts), and is calculated as below.

$\begin{matrix}\left( {{Formula}30} \right) & \end{matrix}$ $\begin{matrix}\left. {{V_{odet}\left( {k_{s} + 1} \right)} = {{\frac{i_{C}\left( {k_{s} + {\left( {m - 1} \right)\frac{T_{sample}}{T_{s}}}} \right)}{C}T_{s}} + {V_{odet}\left( {k_{s} + {\left( {m - 1} \right)\frac{T_{sample}}{T_{s}}}} \right)}}} \right\} & (21)\end{matrix}$

In this regard, Tsample is a time interval for detecting at high speedthe capacitance current iC, and m is the number of times capable ofhigh-speed detection in one control cycle. That is to say, Ts=m−Tsample.

The detected voltage vodet of the output obtained by the abovehigh-speed computing is substituted into the term of vo in Formula 20.Consequently, provided that the command value is iCref, the pulse widthΔT(k) of the manipulated variable can be obtained by the followingFormula 22 using the capacitance current iC alone.

Mode I (Pulse width ΔT(k) in the constant-current discrete control)

$\begin{matrix}\left( {{Formula}31} \right) & \end{matrix}$ $\begin{matrix}{{{\Delta T}(k)} = {\frac{{\frac{L}{3}I_{Cref}} - {\left\{ {\frac{L}{3} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{C - {ave}}\left( k_{s} \right)}}}{V_{in}} + \frac{\left( {T_{s} + T_{d}} \right){v_{odet}\left( k_{s} \right)}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}} & (22)\end{matrix}$

This pulse width ΔT(k) is the manipulated variable for the IC discretecontrol in the mode I.

<Mode II: Buffer Mode During Control Switching>

In contrast to the mode I which employs the constant-current discretecontrol, the mode II and the mode III employ the constant-voltagediscrete control. The constant-voltage discrete control also uses thecapacitance current iC as the detection value, as with theconstant-current discrete control. In order to obtain a controlexpression using the capacitance current iC instead of the inductancecurrent iL, the command value iL(k+1) is defined as below by using again A1.(Formula 32)i _(L)(k+1)=A ₁ {V _(ref) −V _(o)(k _(s))}+i _(R)(k _(s))  (23)

In addition to that, the above Formula 23 is used to modify Formula 11for the pulse width ΔT(k), thereby obtaining the following Formula 24.

Mode II (Pulse Width ΔT(k) in the Constant-Voltage Discrete Control)

$\begin{matrix}\left( {{Formula}33} \right) & \end{matrix}$ $\begin{matrix}{{{\Delta T}(k)} = {\frac{{\frac{L}{3}A_{1}\left\{ {V_{ref} - {V_{o}\left( k_{s} \right)}} \right\}} - {\left\{ {\frac{L}{3} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{C - {ave}}\left( k_{s} \right)}}}{V_{in}} + \frac{\left( {T_{s} + T_{d}} \right){v_{o}\left( k_{s} \right)}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}} & (24)\end{matrix}$

The mode II is a buffer mode to shift from the constant-current controlto the constant-voltage control as shown in FIG. 14 . The buffer modeuses the gain A1 that is smaller than a gain A2 to be used in theconstant-voltage control in the mode III to prevent the occurrence ofthe overshoot and undershoot. Since the output voltage is in transitionin the mode II, the use of a low-speed detected voltage inducessignificantly affect by the delay time. Thus, as with the mode I, theoutput voltage vodet estimated from the capacitance current is used forthe discrete control.

Accordingly, the manipulated variable in the iC discrete control usingthe capacitance current iC in the mode II is expressed by the followingFormulas 25 and 26, in which the gain A1 of the pulse width ΔT(k)expressed by Formula 24 is replaced by gains AH1 and AL1. The gain AH1is a gain on the high power side, and the gain AL1 is a gain on the lowpower side.

Mode II (Pulse width ΔT(k) in the constant-voltage discrete control)

$\begin{matrix}\left( {{Formula}34} \right) & \end{matrix}$ $\begin{matrix}{{{\Delta T}(k)} = {\frac{{\frac{L}{3}A_{H1}\left\{ {V_{ref} - {V_{odet}\left( k_{s} \right)}} \right\}} - {\left\{ {\frac{L}{3} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{C - {ave}}\left( k_{s} \right)}}}{V_{in}} + \frac{\left( {T_{s} + T_{d}} \right){v_{odet}\left( k_{s} \right)}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}} & (25)\end{matrix}$ $\begin{matrix}{{{\Delta T}(k)} = {\frac{{\frac{L}{3}A_{L1}\left\{ {V_{ref} - {V_{odet}\left( k_{s} \right)}} \right\}} - {\left\{ {\frac{L}{3} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{C - {ave}}\left( k_{s} \right)}}}{V_{in}} + \frac{\left( {T_{s} + T_{d}} \right){v_{odet}\left( k_{s} \right)}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}} & (26)\end{matrix}$

<Mode III: Constant-Voltage Discrete Control>

In the mode III, the value iL(k+1) is defined as the command value byFormula 23, as with the mode II. In addition to that, in order toeliminate the affect by the low-speed detected voltage, another gain A2is defined by the following Formula 27.

$\begin{matrix}\left( {{Formula}35} \right) & \end{matrix}$ $\begin{matrix}{A_{2} = \frac{3\left( {T_{s} + T_{d}} \right)}{L}} & (27)\end{matrix}$

With the definition of the gain A2 by Formula 27, the expression of thediscrete control in the mode III can be obtained by the followingFormula 28.

Mode III (Pulse Width ΔT(k) in the Constant-Voltage Discrete Control)

$\begin{matrix}\left( {{Formula}36} \right) & \end{matrix}$ $\begin{matrix}{{{\Delta T}(k)} = {\frac{{\left( {T_{s} + T_{d}} \right)V_{ref}} - {\left\{ {\frac{L}{3} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{C - {ave}}\left( k_{s} \right)}}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}} & (28)\end{matrix}$

Consequently, the term of the low-speed detected voltage vo(ks) isdeleted also in the constant-voltage discrete control, thereby obtainingthe expression of the discrete control using only the capacitancecurrent iC which can be detected by the high-speed detection as thedetection value.

FIG. 15 shows the respective controlling forms, control periods, pulsewidths ΔT(k), command values, output detected voltages, delay times Td,objects to be controlled, gains and others in the mode I, mode II andmode III, and FIG. 16 shows the respective control forms of the mode I,mode II and mode III.

In FIGS. 15 and 16 (a), the mode I conducts the constant-currentdiscrete control in the transition period for shifting between the lowpower side and the high power side, so as to implement theconstant-current discrete control to allow the capacitance current iC ofthe object to be controlled to become the command value IC-ref. In themode I, the detected voltage of the output is estimated from thecapacitance current iC. Furthermore, the output voltage vodet(ks) forthe discrete control at the point ks is estimated from the outputdetected voltage vo(ks−1) at the point ks or the output voltagevodet(ks−1) for the discrete control at the point (ks−1).

In FIGS. 15 and 16 (b), the mode II conducts the constant-voltagediscrete control in the buffer period between the transition period andthe retention period, so as to implement the constant-voltage discretecontrol to allow the detected power voltage vo to be controlled tobecome the command value Vref. In the mode II, the detected voltage ofthe output is also estimated from the capacitance current iC. Theswitching from the mode I to the mode II is conducted after the detectedoutput voltage vo reaches a switching voltage Vc1 or Vc2. The bufferperiod for the control switching by the mode II is one sampling cycle Tsof the control cycle of the control circuit (controller) that switchesthe control to the mode III after one sampling cycle Ts.

In FIGS. 15 and 16 (c), the mode III conducts the constant-voltagediscrete control in the retention period, so as to implement theconstant-voltage discrete control to keep the detected output voltage voto be controlled to the command value vref. In the mode III, the gain A2in the pulse width ΔT(k) is set to 3(Ts+Td)/L to eliminate the term ofthe detected output voltage vo(ks), thereby negating the need for thedetected voltage of the output. By negating the need for the detectedvoltage of the output, the speed of the control can be increased.

Next, a description will be made with reference to FIG. 17 about asignal state in the discrete control by the mode I, mode II and mode IIIin the switching operation of the voltage level of the DC/DC converterof the present invention.

In the following description on the discrete control by each mode,described is an example of the High/Low two-level control for switchingthe power level between the high voltage level (high power side) and thelow voltage level (low power side). The High/Low two-level control isone example, and thus similar discrete control can be employed between aplurality of power levels in which the power levels differ from oneanother.

In the discrete control by the mode I, mode II and mode III, theswitching from the low power side to the high power side and theswitching from the high power side to the low power side are carried outby combining the constant-voltage control and the constant-currentcontrol.

FIG. 17 illustrates the control form of the discrete control by the modeI, mode II and mode III, in which FIG. 17(a) schematically shows thecontrol unit, FIGS. 17(b) and 17(c) respectively show the commandvoltage Vref and the command current IC-ref, respectively, and FIG.17(d) shows the detected output voltage vo. In this figure, thecapacitance current iC is used as the detected current.

<Mode I>

The transition from the low power side to the high power side and thetransition from the high power side to the low poser side are performedby the constant-current discrete control by the mode I. In the mode I,by using the pulse width ΔT(k) obtained by Formula 20 or 22 to retainthe detected current iC to the command current IC-ref, the transition isconducted from the command voltage VL on the low power side toward thecommand voltage VH on the high power side or from the command voltage VHon the high power side toward the command voltage VL on the low powerside.

<Mode II>

At the time that the detected output voltage vo reach the switchingvoltage Vc1 or Vc2, the constant-voltage discrete control is performedto switch from the constant-current control by the mode I to theconstant-voltage control by the mode III. The control period by the modeII is the buffer period for smoothly switching from the constant-currentcontrol to the constant-voltage control. The time width in the bufferperiod can be an integral multiple of the sampling cycle Ts in thecontrol cycle of the control circuit (controller). The time width in thebuffer period is defined to be one sampling cycle Ts to shift from themode I to the mode III in one sampling cycle Ts, thereby increasing thecontrol speed. The time width of the buffer period can be not only onesampling cycle Ts but also n-sampling cycle (n·Ts). In this context, nis an integer.

The mode II uses the pulse width ΔT(k) obtained by Formula 25 to shiftfrom the command voltage VL on the low power side to the command voltageVH on the high power side or from the command voltage VH on the highpower side to the command voltage VL on the low power side. The mode IIis implemented to prevent the overshoot and understood due to theconstant-voltage control. After the constant-voltage discrete control bythe mode II is performed for only one sampling cycle Ts, the control isswitched to the constant-voltage discrete control by the mode III.

<Mode III>

The constant-voltage control by the mode III is conducted after the modeII to control the detected output voltage vo to be the command voltagevalue Vref. In FIG. 17 , the command voltage Vref on the low power sideis defined as VL, and the command voltage Vref on the high power side isdefined as VH.

By implementing the mode I, mode II and mode III, one pulse having twolevels of high and low is formed, and the repetition of these threemodes forms a plurality of pulse outputs. FIGS. 17(b) to 17(d)schematically show the voltage waveforms for purposes of illustration,but not show the actual voltage waveforms.

<Switching of Modes>

In the above-described modes, the transition is repeated in the order ofthe mode I, mode II and mode III.

FIG. 18 is a flowchart showing an example of a mode transition duringshifting from the low power side to the high power side. Although thisexample shows that the mode II is carried out in a cycle of the onesampling cycle, the mode II may be carried out in several cycles.Moreover, these cycles are not limited to the sampling cycle, and may bean arbitrary cycle.

In response to a command for shifting from the low power side to thehigh power side (s1), the detected output voltage vo(k) obtained by thecomputation at the control sampling time after the command is calculatedas vodet(ks), and the voltage detection value at the point of samplingis defined as a computation default value vodet(0) for computing theoutput voltage vodet(k),

In response to a command for shifting from the high power side to thelow power side (s1), the detection value vo(k) of the output voltageobtained at the point right before the shifting is defined as a defaultvalue vo(ks) (s2), and the computation is conducted in the mode I at thecontrol sampling time after the control cycle Ts by using Formula 19 toestimate a value vodet(ks+1) as the output voltage (s3). As to the valuevodet after the point (ks+1), the values vodet(ks+1), vodet(ks+2) andvoldet (ks+3) can also be estimated in a similar way.

In response to a command for shifting from the high power side to thelow power side (s1), the detection value vo(k) of the output voltageobtained at the point right before the shifting is defined as a defaultvalue vo(ks) (s2), and the computation is conducted in the mode I at thecontrol sampling time after the control cycle Ts by using Formula 19 toestimate a value vodet(ks+1) as the output voltage (s3). As to the valuevodet after the point (ks+1), the values vodet(ks+1), vodet(ks+2) andvoldet (ks+3) can also be estimated in a similar way.

The shift command is switched from the high power side to the low powerside (s7), and then the shift is carried out in similar way to theabove-described steps s2 to s6. This shift is from the high power sideto the low power side.

<Switching Voltage Vc1, Vc2>

The switching voltages Vc1 and Vc2 from the mode I to the mode II arecalculated by the following Formulas 29 and 30, respectively.

$\begin{matrix}\left( {{Formula}37} \right) & \end{matrix}$ $\begin{matrix}{V_{c1} = {V_{Href} - {\frac{{3T_{s}} + {2T_{d}}}{2C_{o}}I_{cref}}}} & (29)\end{matrix}$ $\begin{matrix}\left( {{Formula}38} \right) & \end{matrix}$ $\begin{matrix}{V_{c2} = {V_{Lref} - {\frac{{3T_{s}} + {2T_{d}}}{2C_{o}}I_{cref}}}} & (30)\end{matrix}$

The voltage Vc1 is the switching voltage used for switching from the lowpower side to the high power side, and the voltage Vc2 is the switchingvoltage used for switching from the high power side to the low powerside.

The switching voltages Vc1, Vc2 are defined by taking into considerationthe maximum change of voltage at an interval to switch to the mode III,and is a value of generation limit voltage in which the change in thevoltage at the starting time of the mode III leads to the overshoot orundershoot. For example, the change in the voltage at the maximum timeTs caused by jitter of the switching voltage, the change in the voltageoccurring in one sample after altering the command value, and the changein the voltage during the control delay time Td are taken into accountto select the voltage that does not cause the overshoot.

The switching voltages Vc1, Vc2 are for implementing the mode switchingby subtracting from the voltage command value the maximum change in thevoltage during the time until switching to the mode III to therebypreventing the overshoot in all conditions.

That is to say, the change in the voltage at a maximum time T_(s) causedby the jitter of the switching voltage ((Ts/Co)·ICref), the change inthe voltage occurring in one sample after the command value is changed((Ts/2Co)·ICref), and the change in the voltage during the control delaytime Td ((Td/Co)·ICref) are subtracted from the voltage command valueVHref to obtain Formulas 29 and 30. In this context, Co is the outputcapacity of the main circuit.

The switching voltages Vc1, Vc2 respectively expressed by Formulas 29,30 are examples of the three-phase interleaving that takes intoconsideration the maximum change in the voltage, and thus if the numberof phase is n, the coefficient of Ts in the formula can be replaced by nfrom three, or if the maximum change in the voltage is acceptable, theswitching voltages Vc1, Vc2 can be multiplied by a coefficient having apredetermined value smaller than 1.

<Gain A1 (AH1, AL1)>

The manipulated variables for the discrete control in the mode II areexpressed by Formulas (25) and (26), respectively, and a gain A1 (AH1,AL1) included in each formula prevents the overshoot and undershoot.Now, the range of the gain A1 (AH1, AL1) in the mode II will bedescribed. In this regard, the description refers to the transition fromthe low power side to the high power side.

If the voltage command value in the high power is defined as VHref, thevoltage detection value computed based on the capacitance current iC inthe mode II is defined as Vodet-mode2 and the detected current value inthe mode III is defined as Vo-mode3, the capacitance current commandvalue in each mode is represented as below.

Mode I: ICref

Mode II: AH1(VHref−Vodet−mode2)

Mode III: A2(VHref−Vo−mode3)≈0, wherein A2=3(Ts+Td)/L

Since the mode II is the buffer period between the mode I and the modeIII, the capacitance current command in the mode II is in the rangebetween the mode I and the mode III, and thus has the relationship ofICref>AH1(VHref−Vodet−mode2)>0.

By using the above magnitude relationship and an evaluation voltage VC1for switching expressed by Formula 29, the range of the gain AH1 isexpressed by the following formula.

$\begin{matrix}\left( {{Formula}39} \right) & \end{matrix}$ $\begin{matrix}{0 < A_{H1} < \frac{2C_{o}}{T_{s} + {2T_{d}}}} & (31)\end{matrix}$

Thus, the gain AH1 is used as a coefficient for determining followingcharacteristic for the command voltage VHref on the high side in theaforementioned range. The gain AL1 will not be described in here, butcan be processed in the same way as the gain AH1.

<Gain A2>

In Formula 24 expressing the pulse width T(k) of the constant-voltagediscrete control, the term of the detected voltage is{(L/3)·A1·vo(ks)/Vin} and {(Td+Ts)·vo(ks)/Vin}. By applying Formula 24to the mode III to replace the gain A1 by the gain A2 and defining thegain A2 by Formula 27, the terms of two detected voltages are balancedout each other, thereby removing the term of vo(k) of the outputvoltage. Consequently, the expression of the discrete control in themode III is expressed by Formula 28 that does not include the detectedoutput voltage vo(k).

Concerning the pulse widths ΔT(k) in each of the modes I, II and III,each formula (shown as “High”) for the transition from the low powerside to the high power side and each formula(shown as “Low”) for thetransition from the high power side to the low power side arecollectively presented in the following formula.

$\begin{matrix}{{High}\left\{ {\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}{mode1} \\{{{If}v_{o}{\det\left( k_{s} \right)}\left( {{Formula}(21)} \right)} \leq {v_{c1}\left( {{Formula}(29)} \right)}}\end{matrix} \\{{{Then}{{\Delta T}(k)}} = {\frac{{\frac{L}{3}f_{Cref}} - {\left\{ {\frac{L}{3} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{C - {ave}}\left( k_{s} \right)}}}{V_{in}} +}}\end{matrix} \\{\frac{\left( {T_{s} + T_{d}} \right){v_{odet}\left( k_{s} \right)}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}\end{matrix} \\{mode2}\end{matrix} \\{{{If}v_{o}{\det({ks})}} > {v_{c1}{then}{the}{operation}{period}{is}1{cicle}T{at}{AH}_{1}}}\end{matrix} \\{{{\Delta T}(k)} = {\frac{{\frac{L}{3}A_{H1}\left\{ {V_{Href} - {V_{odet}\left( k_{s} \right)}} \right\}} - {\left\{ {\frac{L}{3} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{C - {ave}}\left( k_{s} \right)}}}{V_{in}} +}}\end{matrix} \\{\frac{\left( {T_{s} + T_{d}} \right){v_{odet}\left( k_{s} \right)}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}\end{matrix} \\{mode3}\end{matrix} \\{{The}{next}{operation}{is}{}{{AH}_{2}\left( {A_{v} = {3{T_{s}/L}}} \right)}}\end{matrix} \\{{{\Delta T}(k)} = {\frac{{\left( {T_{s} + T_{d}} \right)V_{Href}} - {\left\{ {\frac{L}{3} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{C - {ave}}\left( k_{s} \right)}}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}}\end{matrix}{L{ow}}\left\{ \begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}\begin{matrix}{mode1} \\{{{If}v_{o}{\det\left( k_{s} \right)}\left( {{Formula}(21)} \right)} \geq {v_{c1}\left( {{Formula}(30)} \right)}}\end{matrix} \\{{{Then}{{\Delta T}(k)}} = {\frac{{{- \frac{L}{3}}I_{Cref}} - {\left\{ {\frac{L}{3} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{C - {ave}}\left( k_{s} \right)}}}{V_{in}} +}}\end{matrix} \\{\frac{\left( {T_{s} + T_{d}} \right){v_{odet}\left( k_{s} \right)}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}\end{matrix} \\{mode2}\end{matrix} \\{{{If}v_{o}{\det({ks})}} > {v_{c2}{then}{the}{operation}{period}{is}1{cicle}T{at}{AH}_{1}}}\end{matrix} \\{{{\Delta T}(k)} = {\frac{{\frac{L}{3}A_{H1}\left\{ {V_{Lref} - {V_{odet}\left( k_{s} \right)}} \right\}} - {\left\{ {\frac{L}{3} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{C - {ave}}\left( k_{s} \right)}}}{V_{in}} +}}\end{matrix} \\{\frac{\left( {T_{s} + T_{d}} \right){v_{odet}\left( k_{s} \right)}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}\end{matrix} \\{mode3}\end{matrix} \\{{The}{next}{operation}{is}{}{{AH}_{2}\left( {A_{v} = {3{T_{s}/L}}} \right)}}\end{matrix} \\{{{\Delta T}(k)} = {\frac{{\left( {T_{s} + T_{d}} \right)V_{Lref}} - {\left\{ {\frac{L}{3} - \frac{\left( {T_{s} + T_{d}} \right)^{2}}{2C}} \right\}{i_{C - {ave}}\left( k_{s} \right)}}}{V_{in}} - {\frac{T_{d}}{T_{s}}{{\Delta T}\left( {k - 1} \right)}}}}\end{matrix} \right.} \right.} & \left( {{Formula}40} \right)\end{matrix}$

Furthermore, there is a delay about 1 us even in a conventionalhigh-speed DC current sensor for detecting an inductance current. Bycontrast, there are many AC current sensors with response capability of10 MHz or more (0.1 us or less delay). Thus, the capacitance currentwhich can be detected by the AC current sensor is used for performingthe IC discrete control with the capacitance current iC to attain thehigh-speed control. It is to be noted that the above numerical value ofthe response capability of the sensor is an example which is not limitedthereto, and the AC current sensor typically have the higher responsecapability than that of the DC current sensors.

In a verification by employing a circuit simulation and using actualequipment, the behavior of the High/Low pulse operation by the DC/DCconverter of the present invention was measured, and thereby theeffectiveness of the discrete control considering the control delay hasbeen confirmed.

Examples of Application to DC Power Supply Device and AC Power SupplyDevice

Next, a description will be made on the examples of application of theDC/DC converter of the present invention to a DC power supply device andan AC power supply device, by referring to FIG. 19 .

FIG. 19 is a diagram of control block illustrating the control system inthe examples of application of the DC/DC converter of the presentinvention to the DC power supply device and the AC power supply device.

The control system of the control blocks shown in FIG. 19(a) is anexample of the configuration including PI control constituting a mainloop control system and discrete control constituting a minor loopcontrol system, and the control system of the control block shown inFIG. 19(b) is an example of the configuration including only thediscrete control constituting the minor loop control system.

The configuration shown in FIG. 19(a) forms the command voltages VH, VLby the PI control on the basis of the command powers PH, PL in the mainloop control system, so as to conduct the discrete control in the minorloop control system.

In addition to that, the configuration shown in FIG. 19(b) conducts thediscrete control in the minor loop control system based on the givencommand voltages VH, VL. If the command voltages VH, VL are obtained,the discrete control can be implemented without the need for the mainloop control system.

With respect to the discrete control constituting the minor loop controlsystem, the present invention applies the two-level discrete controlsystem that performs the control according to DC command voltages of thehigh level command voltage VH and the low level command voltage VL inthe bilateral step-down chopper circuit with the multi-phaseinterleaving system of the DC/DC converter of the present invention.

When the two-level control in the high level and the low level isperformed, the high level power command PH and the low level powercommand PL are used as command signals in the main loop to conduct thePI control by detecting the power obtained from the load side, therebyobtaining the high level command voltage VH and the low level commandvoltage VL.

In the minor loop, the high level command voltage VH and the low levelcommand voltage VL obtained by the PI control are used as the commandvalues, so that the detected output voltage vo or the capacitancecurrent iC are detected to implement the discrete control.

As the descriptions about the embodiments and their variations arepresented as examples of the DC/DC converter in accordance with thepresent invention and are not limited thereto, the variation can beimplemented in many ways based on the purpose of the present invention,and thus these variations are not excluded from the scope of the presentinvention.

INDUSTRIAL APPLICABILITY

The DC/DC converter of the present invention is applicable to equipmentfor manufacturing semiconductors, liquid crystal display panels andothers, vacuum deposition equipment, and applicable for supplyinghigh-frequency power to an apparatus that uses a high-frequency wave,such as heating and fusion apparatus.

REFERENCE SIGNS LIST

-   1 DC/DC Converter-   2 Main Circuit (Chopper Circuit)-   3 Switching Circuit-   4 LC Circuit-   5 Switching Signal Generator-   6 Control Unit-   7 Load

The invention claimed is:
 1. A DC/DC converter, comprising: a maincircuit including a switching circuit; a control unit for performingdiscrete control toward a command value; and a switching signalgenerator for generating a switching signal, wherein the control unitcomputing a pulse width ΔT(k) of a switching signal having a delay timeTd as a parameter for a manipulated variable of the discrete controlconducted for every control cycle Ts, the switching signal generatorgenerating the switching signal based on the pulse width ΔT(k), the maincircuit driving the switching circuit based on the switching signal,wherein, the main circuit comprises the switching circuit and an LCcircuit composed of a series-parallel circuit of an inductance LA and acapacitance C, in which pulse widths ΔT(k) of a cycle (k) of the maincircuit in the discrete control by the control unit has a voltagecomponent across the capacitance C, and an input voltage vo in the LCcircuit, and a pulse width ΔT(k−1) of a cycle (k−1) of the main circuit,the voltage component and the input voltage have an additional value(Ts+Td) as a parameter, which is obtained by adding the delay time Td tothe control cycle Ts, and the pulse width ΔT(k−1) has a coefficient of(Td/Ts) obtained by dividing the delay time Td by the control cycle Ts.2. The DC/DC converter according to claim 1, wherein the discretecontrol is a single-phase or multi-phase interleaving control, in whichthe inductance value of the pulse width ΔT(k) in the multi-phaseinterleaving control of n-phase (wherein n is an integer of 2 or more)is 1/n of the inductance value of the pulse width ΔT(k) in thesingle-phase interleaving control.
 3. A control method for a DC/DCconverter that comprises: a main circuit including a switching circuit;a control unit for performing discrete control toward a command value;and a switching signal generator for generating a switching signal,wherein the control unit computing a pulse width ΔT(k) of a switchingsignal having a delay time Td as a parameter for a manipulated variableof the discrete control conducted for every control cycle Ts, theswitching signal generator generating the switching signal based on thepulse width ΔT(k), the main circuit driving the switching circuit basedon the switching signal, wherein, the main circuit comprises theswitching circuit and an LC circuit composed of a series-parallelcircuit of an inductance LA and a capacitance C, in which pulse widthsΔT(k) of a cycle (k) of the main circuit in the discrete control by thecontrol unit has a voltage component across the capacitance C, and aninput voltage vo in the LC circuit, and a pulse width ΔT(k−1) of a cycle(k−1) of the main circuit, the voltage component and the input voltagehave an additional value (Ts+Td) as a parameter, which is obtained byadding the delay time Td to the control cycle Ts, and the pulse widthΔT(k−1) has a coefficient of (Td/Ts) obtained by dividing the delay timeTd by the control cycle Ts.
 4. The control method for the DC/DCconverter according to claim 3, wherein the discrete control is asingle-phase or multi-phase interleaving control.